Bipolar junction transistor structure

ABSTRACT

We disclose a bi-directional bipolar junction transistor (BJT) structure, comprising: a base region of a first conductivity type, wherein said base region constitutes a drift region of said structure; first and second collector/emitter (CE) regions, each of a second conductivity type adjacent opposite ends of said base region; wherein said base region is lightly doped relative to said collector/emitter regions; the structure further comprising: a base connection to said base region, wherein said base connection is within or adjacent to said first collector/emitter region.

FIELD OF THE INVENTION

This invention relates to a design of semiconductor power transistor,particularly but not exclusively, to a semiconductor power switch andrelays.

BACKGROUND OF THE INVENTION

Traditional power switches are fabricated from P and N type regions ofsemiconductor material with metalised layers and sometimes with thinoxide layers for field-effect devices. Well known in the art aretechnologies such as Bipolar Junction Transistors (BJT), Junction FieldEffect Transistors (JFET), Metal Oxide Field Effect Transistors (MOSFET)and Insulated Gate Biploar Transistor (IGBT) which generally switch DCvoltages. Whereas Triacs, Silicon Controlled Rectifiers (SCR) as well asmodified (typically back-to-back) versions of the DC switch types havebeen used to switch AC power. Whether fabricated on Silicon or acompound semiconductor such has Silicon Carbide (SiC) all these standarddevices has one or more limitation when it comes to the common task ofswitching AC power on and off.

These limitations lead to the following problems.

-   -   Lack of AC switching ability.    -   Expensive semiconductor materials.    -   Complex semiconductor processing steps.    -   Inherent voltage drop on turn on (typ. 0.8 to 2.5 volts)        limiting efficiency.    -   Lack of turn-off ability—can prevent implementation of        short-circuit protection feature.    -   Complex, high-voltage, high current +ve and −ve base/gate drive        circuitry.    -   High on-resistance for high voltage devices.

It is an object of the present invention to address the problemsdescribed above.

SUMMARY OF THE INVENTION

The proposed new device can be manufactured at low cost on standard BJTfabrication equipment, has inherent AC switching ability and simpledrive requirements. Voltage drop at turn on can be under 100 mV at 10'sof amps per cm² and the device can switch high voltages both on and offat will with a single polarity drive pulse without latch-up. Inconjunction with a low cost microcontroller, these switches can performefficient mains voltage switching, short circuit protection, loaddiagnostics and data logging functions for smart-power, smart appliancesystems without requiring special heatsinks or fan cooling.

According to the present invention there is therefore provided abi-directional bipolar junction transistor (BJT) structure, comprising:a base region of a first conductivity type, wherein said base regionconstitutes a drift region of said structure; first and secondcollector/emitter (CE) regions, each of a second conductivity typeadjacent opposite ends of said base region; wherein said base region islightly doped relative to said collector/emitter regions; the structurefurther comprising: base connection to said base region, wherein saidbase connection is within or adjacent to said first collector/emitterregion.

It will be appreciated that the term “drift region” refers to a highvoltage sustaining region including a relatively low dopingconcentration. During the off-state of the device, the drift region issubstantially fully depleted. In a conventional high voltage BJT, thecollector region generally acts as the high voltage sustaining layer. Onthe contrary, the thick base, drift region works as the high voltagesustaining layer. Furthermore, in the conventional high voltageThyristor, the main current conduction generally takes place throughfour active semiconductor layers, e.g. cathode (N), gate (P), ndrift (N)and anode (P). By contrast, the bi-directional device of the presentinvention provides current conduction through three active layers, i.e.first CE region, base region and the second CE region.

Broadly speaking the base connection and base region define a secondtransistor which enables the bi-directional BJT structure to support ahigh voltage in a case where, in effect, a base emitter junction of thestructure is forward biased so that a high current through this junctionwould otherwise flow. This second transistor may either constitute afurther BJT or may be a JFET, depending upon whether the base connectionis within or in the vicinity of/adjacent to the first CE region,respectively. Thus where the base connection is within the first CEregion the second transistor is a BJT transistor with collector/emitterterminals connected to the base connection and to the base, drift regionrespectively and with its base connection formed by the first CE regionof the bi-directional BJT structure. Typically the first and second CEregions of the bi-directional BJT structure are heavily doped, forexample in a range between 10¹⁸ cm⁻³ to 10²¹ cm⁻³ whilst the base regionis lightly doped compared to the CE regions. The first CE region mayextend laterally to form the base of the second transistor. Thus, forexample, the first and second CE regions of the BJT structure may be N⁺type and the base region of the bi-directional BJT structure P⁻, and thesecond transistor may then be a PNP transistor with the base regioncomprising a preferably relatively lower doped, N⁻ extension of thefirst, N⁺ CE region, and with collector/emitter terminals of the secondtransistor comprising the P⁻ base region and the base connection. Inpreferred embodiments the base connection is an ohmic connectioncomprising a region of the first conductivity type, typically heavilydoped, for example a range between 10¹⁸ cm⁻³ to 10²¹ cm⁻, such as a P⁺region.

In other embodiments the second transistor is a JFET and the baseconnection/ohmic contact (P+) and base, drift region constitutesource/drain connections of the JFET. The gate terminal of the JFET isthen formed by the (heavily doped, N⁺) first CE region of thebi-directional BJT structure. In this case the (ohmic) base connectionis adjacent (which here includes slightly spaced apart from) the firstCE region. The channel region of the JFET lies between the (ohmic) baseconnection and the base drift region so that, in effect, the JFETcontrols conduction between the base connection and the base driftregion.

It can therefore be appreciated that where there is a forward conductionpath from the base region to the second CE region the second transistoreffectively stops substantially the entire voltage of what may be a highvoltage across the device appearing between the base connection and thesecond CE region—this would otherwise drive a large current through thedevice which would destroy it. Instead when a forward conduction pathbetween the base region and the second CE region is present (driven by avoltage on the base region relative to the second CE region) a forwardconduction path between the base connection and the base region includesa depleted portion of the base region—that is the base connection of thestructure is effectively isolated by the depleted portion of the baseregion. Thus during switch on, when a high voltage can appear across thefirst and second CE regions, substantially all this voltage can alsoappear between the base connection and second CE region across the basedepletion region. As the device turns on the voltage across the firstand second CE terminals will fall to a very low voltage and this ceasesto be a potential problem.

Preferably when the base region and the base connection are of p-typeand the first and second CE regions are of n-type, the following on andoff states occur in the BJT structure: (1) when no voltage is applied toany terminals, the structure may be in an off-state so as to formdepletion regions between said first CE region and base region andbetween said second CE region and base region; (2) when a positivevoltage is applied to said second CE region and no voltage is applied tothe first CE region and the base connection, the structure is in anoff-state so as to form a depletion region between said second CE regionand base region; (3) when a negative voltage is applied to said secondCE region and no voltage is applied to the first CE region and the baseconnection, the structure may be in an off-state so as to form adepletion region between said first CE region and base region; (4) whena first positive voltage is applied to said second CE region, a secondpositive voltage being applied to the base connection and no voltage isapplied to the first CE region, the structure is in an on-state in whichmajority carriers from the first CE region flow through the base regiontowards the second CE region, and minority carriers from the baseconnection are injected into the base region, the minority carriersbeing recombined with the majority carriers in a region adjacent thefirst CE region; (5) when a negative voltage is applied to said secondCE region, a positive voltage being applied to the base connection andno voltage is applied to the first CE region, the structure is in anon-state in which majority carriers from the second CE region flowthrough the base region towards the first CE region, and minoritycarriers from the base connection are injected into the base regionflowing towards the second CE region, the minority carriers beingrecombined with the majority carriers in a region adjacent the second CEregion. Here the majority carriers are electrons and the minoritycarriers are holes.

Preferably when the base region and the base connection are of n-typeand the first and second CE regions are of p-type, the following on andoff states occur in the BJT structure: (1) when no voltage is applied toany terminals, the structure may be in an off-state so as to formdepletion regions between said first CE region and base region andbetween said second CE region and base region; (2) when a positivevoltage is applied to said second CE region and no voltage is applied tothe first CE region and the base connection, the structure is in anoff-state so as to form a depletion region between said first CE regionand base region; (3) when a negative voltage is applied to said secondCE region and no voltage is applied to the first CE region and the baseconnection, the structure may be in an off-state so as to form adepletion region between said second CE region and base region; (4) whena first positive voltage is applied to said second CE region, a negativevoltage being applied to the base connection and no voltage is appliedto the first CE region, the structure is in an on-state in which holesfrom the second CE region flow through the base region towards thesecond CE region, and electron from the base connection are injectedinto the base region, the electrons being recombined with the holes in aregion adjacent the second CE region; (5) when a negative voltage isapplied to said second CE region, a negative voltage being applied tothe base connection and no voltage is applied to the first CE region,the structure is in an on-state in which holes from the first CE regionflow through the base region towards the second CE region, and electronsfrom the base connection are injected into the base region flowingtowards the first CE region, the electrons being recombined with theholes in a region adjacent the first CE region.

Conveniently, in some preferred embodiments the base connection isrecessed into a surface of the structure—conveniently an ohmicconnection of the first conductivity type can be formed within arecessed portion of the first CE region. Conveniently the recess may besufficient to also incorporate a metal connection to the ohmic forexample P⁺ region. It will be appreciated that in the proposed devicethe base region may be ohmic in nature so as to drive the maintransistor (not the second transistor) comprising the first CE region,base region and second CE region into the saturation region in whichboth saturation diffusion and drift current apply.

It will be appreciated that the labelling of the first and second CEregions is arbitrary. Although In some preferred embodiments the deviceis a vertical device, a lateral device may also be fabricated. Forexample the first and second CE regions may be fabricated in a commonlayer, displaced laterally from one another, separated by a base regionwhich runs beneath the first and second CE regions and laterally,joining these regions to one another.

In some preferred embodiments the base, drift region is wider in adirection between the ends of the region adjacent the CE regions thaneach CE region. However in some very high voltage devices the CE regionsthemselves may be relatively wide, comprising a long (deep) diffusion.In embodiments a current carrying capability of a connection pathbetween the base connection and the second CE region is less than acurrent carrying capability of a connection path between the first andsecond CE regions—that is the main conduction path through the device isbetween the first and second CE regions, the base connection merelyproviding a relatively small base current. It will further beappreciated that the structure is non-latching, that is a connectionbetween the first and second CE regions is switched off on removal of avoltage from the base connection (without requiring a current betweenthe first and second CE regions to go to zero).

According to a further aspect of the present invention there is provideda bipolar junction transistor (BJT) structure, comprising: a base regionof a first conductivity type, wherein said base region constitutes adrift region of said structure; first and second collector/emitter (CE)regions, each of a second conductivity type adjacent opposite ends ofsaid base region; wherein said base region is lightly doped relative tosaid collector/emitter regions; the structure further comprising: a baseconnection to said base region, wherein said base connection is withinor adjacent to said first collector/emitter region and a buried layer ofthe second conductivity type disposed between the second CE region andthe base region.

This bipolar junction transistor structure could be termed as anon-insulated gated bipolar junction transistor (NIGBT). It will beappreciated that the NIGBT devices preferably operate in DC applicationsin which the buried layer can help to sustain high voltage due to thepunch-through structure. The NIGBT structure is capable of operating inAC mode as well. However, in the AC mode, voltage sustaining capabilitycan be limited in one direction (or during a DC application) as thedevice try to deplete the highly doped buried layer.

According to a further aspect of the present invention there is provideda bipolar junction transistor (BJT) structure, comprising: a base regionof a first conductivity type, wherein said base region constitutes adrift region of said structure, the drift region being a reverse voltagesustaining region; an emitter region of a second conductivity type; acollector of a second conductivity type, the collector and emitter beingadjacent opposite ends of said base region; wherein said base region islightly doped relative to said collector and emitter regions; thestructure further comprising: a base connection region of the firstconductivity type formed adjacent to said emitter region and a fieldstop layer of the first conductivity type formed between the emitterregion and the base region, the base connection being within the fieldstop layer.

The doping concentration of the field stop layer may be less than thatof the base connection. The thickness of the field stop layer may bemore than that of the base connection. The BJT structure may beconfigured such that a diode is formed between the collector and baseregion. The diode may be configured to operate as a reverse conductingdiode when driven by a driver circuit. The driver circuit may include asoftware controlled driver which can allow the BJT structure to provethe free-wheeling diode characteristics. The software controlled drivermay be configured such that the free-wheeling virtual diode may notallow large reverse voltage to occur.

The invention may provide a driver circuit operatively connected to theBJT structures above, the driver circuit comprising a first PWMcontroller and a second PWM controller, the first and second PWMcontrollers being coupled to one another, wherein the first PWMcontroller is capable of controlling a low voltage transistor and thesecond PWM controller is a high-frequency converter. The invention mayalso provide a driver circuit operatively connected to a standard BJTstructure—thus it would be apparent that the driver circuit described inthis specification are not limited to the proposed BJT structuresdisclosed but they are applicable to standard BJT structures as well.

The second PWM controller may be a buck converter using an inductor. Thecross modulation of the first and second PWM controllers across a basedrive inductance may allow a continuous dynamic current control of theBJT structure with high speed on and off capability. The frequency ofthe second PWM controller may be such that the off time is ofsubstantially similar order compared to the minority carrier (e.g.electron) lifetime ensuring that conductivity of the BJT structureremains substantially constant during the period controlled by thesecond PWM controller. The second PWM controller may be a phase offsetcontroller. The driver circuit may further comprise a third PWMcontroller which is a phase offset controller, wherein the second andthird phase offset controllers each drive an inductor with a commonpoint on a base connection terminal of a transistor. The first PWMcontroller may be coupled to the base connection terminal of thetransistor so as to drive the transistor. The multiphase offsets of thesecond and third PWM controllers create a relatively high effectivefrequency to match a reduced minority-carrier lifetime of high speedtransistors. The combination of the driver circuit and the BJT structuremay be configured to provide a reverse conducting diode.

When the base region is of p conductivity type, the driver may beconfigured to detect a negative current of the base so that holes can bepulled out from the junction between the first CE region and base toclamp a negative excursion.

The current rating of the base region substantially equal to that of thecollector or emitter so as to yield a reverse free-wheeling diode ofsubstantially equal current rating to the forward rating.

The base region may be configured to be clamped to ground using a MOSFETfrom the driver. The base current may be produced by the driver circuitto result in a reverse bias switch-on transistor action so as to reducethe voltage drop in a current flow path below a normal voltage drop of adiode.

A computer program product comprising a computer readable medium inwhich a computer program is stored, the computer program comprisingcomputer readable code which, when run by a controller of a drivercircuit, causes the driver circuit to operate as the driver circuitabove, wherein the computer program is stored on the computer readablemedium.

When the controller is configured to detect a voltage of the BJTstructure in an off-state, the computer program product may configuredto turn on the BJT structure to emulate a reverse conducting diodeaction.

In a related aspect the invention provides a driver circuit, inparticular for a structure as described above, comprising a voltagesensing resistor and a current sensing resistor each coupled to one ofthe CE regions (optionally the same region). A microcontroller iscoupled to the resistors to receive respective voltage sensing andcurrent sensing signals. In embodiments the microcontroller isconfigured to provide a PWM (pulse width modulation) output forcontrolling a current into the base connection of the structure, via aninductance. This arrangement facilitates accurate control of the devicealong a defined operating path. The voltage across the device andcurrent through the device are sensed and the current into the baseconnection may then be adjusted to move the device between a switched-onand a switched-off configuration in a controlled manner.

In a further aspect the invention also provides a circuit breakercomprising a first, power semiconductor switching device and a drivercircuit, wherein said circuit breaker has two power switching terminals,and further comprises a power supply, and a controller for said powersemiconductor switching device powered by said power supply, whereinsaid power supply is coupled in series with said first powersemiconductor switching device between said power switching terminals toderive a power supply from said terminals whilst said powersemiconductor switching device is on, and wherein said power supplycomprises a second switching device coupled in series with said firstpower semiconductor switching device, between said power switchingterminals, such that the circuit breaker is operable without a separatepower supply.

In embodiments, by employing a second switching device in series withthe first the relatively high current through the power switching devicewhen on can be leveraged with only a very small voltage drop to generatesufficient power for driving a base current into the device. It will beappreciated that when the power switching device is on the voltage dropacross the two power switching terminals should be as low aspracticable, and by employing a second, low voltage switching device inseries with the first a reasonable power, for example of order 1 wattcan be achieved with a very small voltage drop, for example of order 0.1volts, without wastage in, for example, a lead resistor. In preferredembodiments the power switching device is a high voltage device and thesecond device is a low voltage device, in particular forming part of theinput stage of a DC-to-DC converter. In this context a power devicerefers to a high voltage device which typically operates with a voltagein the range greater than 100 volts, 500 volts or 1000 volts and or atcurrents at greater than 1 amp, 10 amps or 100 amps. A low voltagedevice typically operates at a voltage of less than 50 volts, inembodiments less than 10 volts.

It is preferable that the circuit breaker can be installed either wayaround in a circuit and thus the direction of current flow between thepower switching terminals may not be known. In preferred embodiments,therefore, the power supply is a switched mode power convertercomprising a plurality of low voltage switching devices arranged tocharge and discharge an energy storage component, (capacitor and/orinductor) so that power for the controller is provided with the sameplurality no matter which way round the circuit breaker is connectedinto each circuit. Thus the power supply may further comprise a sensorto sense a direction of current flow for controlling the plurality ofswitching devices according. Broadly speaking the switches are arrangedso that whichever the direction the current flows a positive side of theenergy storage component delivers power to a positive line for thecontroller, and vice versa. This can be achieved by sensing thedirection of current flow through the circuit breaker in order todetermine which of the two terminals is positive with respect to theother, so that the switches can be controlled accordingly. Preferablythe power supply also includes an arrangement to ensure proper start-upof the circuit breaker. In embodiments this comprises a reservoircapacitor charged by leakage current through the power switching devicewhen the power switching device is off. In embodiments the power supplyfrom this leakage current is sufficient to operate a microcontroller orother circuit to sense the direction of current flow through the circuitbreaker, i.e. the orientation in which a circuit breaker is connected,before the power switching device has switched on, and thus when thedevice switches on can automatically start up the switched mode powerconverter to provide power of the correct polarity to the controller.

The invention further provides a circuit breaker operably connected tothe BJT structure above, the circuit breaker comprising: an inputcapacitor connected to a CE region; an inductor coupled to the inputcapacitor; first and second switching devices coupled to the inductor; asecond capacitor coupled to the second switching device; and a pulsewidth modulation (PWM) controller configured to control the first andsecond switching devices. It will be appreciated that the circuitbreaker can be operably connected to a standard BJT structure.

When a positive voltage is applied to the CE terminal, the firstswitching device may be configured to charge the inductor, and thesecond switching device may be configured to charge the secondcapacitor. The charging of the inductor may be controlled by controllingthe duty cycles of the PWM controller. When a negative voltage isapplied to the first CE terminal, the second switching device and thesecond capacitor may be disconnected from the circuit breaker.

The circuit breaker above may further comprise a third switching deviceand a third capacitor which are coupled to the first switching device,the inductor and the first capacitor. The third switch may be configuredto charge the third capacitor.

The invention may further provide a bootstrap circuit operativelyconnected to the BJT structure above and operatively connected to thecircuit breaker above, the bootstrap circuit comprising a first diodecoupled with the second capacitor of the circuit breaker and a seconddiode coupled with the third capacitor of the circuit breaker, whereinthe bootstrap circuit is configured to store positive or negativeleakage current in the first and/or third capacitors through the firstand second diodes so as to turn on the bi-direction BJT structure. Itwill be appreciated that the bootstrap circuit breaker can be operablyconnected to a standard BJT structure.

The bootstrap circuit may further comprise a bleed resistor to providesufficient current to turn on the BJT structure if there is inherentleakage current present in the BJT structure.

The bootstrap circuit may further comprise an auxiliary tap circuitswitching on around the zero-crossing times so as to power the BJTstructure.

The invention may further provide a driver circuit operatively connectedto a plurality of BJT structures above, wherein each BJT structure isdisposed side by side on a chip and wherein the driver circuit comprisesa plurality of independent PWM drivers each independently driving thebase connection of each BJT structure through an inductor. Each PWMdriver may be configured to control current to the base connection andswitching time of the BJT structure independently. It will beappreciated that the driver circuit can be operably connected to aplurality of standard BJT structures.

Each PWM driver may be configured to control the current during anon-state of the BJT structure using a discontinuous current inductordrive.

The discontinuous current mode may occur when an off-time from the PWMdriver is sufficiently long so that the inductor current decreases tozero.

The invention may further provide a driver circuit operatively connectedto a BJT structures above or to a standard BJT structure, the drivercircuit comprising a resistive digital to analogue controller (DAC) forcontrolling the current of the base of the BJT structure. The DAC may beconfigured to control the base current of the BJT structure according toa control program which is reactive to measured operating conditions ofthe BJT structure.

The invention may further provide a matrix converter comprising an arrayof BJT structures above, the matrix converter further comprising acontrol circuit comprising a plurality of channels which are configuredto control the switching of the array of BJT structures.

The invention may further comprise a relay circuit for a low leakagecurrent application, the relay circuit comprising the BJT structureabove or a standard BJT structure, the relay circuit further comprisinga load resistor and a switching device arranged parallel to the loadresistor, wherein the switching device is configured to bypass anyleakage current from the BJT structure around the load resistor duringswitching off operation.

The relay circuit may further comprise a further switching devicecoupled with the load resistor, the further switching device beingconfigured to obtain Pico-ampere level leakage current into the loadresistor.

The invention may further provide a driver chip operatively connected toa BJT structure above or to a standard BJT structure and may comprisethe driver circuit above, wherein the driver chip is configured to applypre-programmed coefficients determined after manufacturing thecomponents of the driver chip.

The first PWM controller may be configured to vary phases for differentregions of the BJT structure based on calibration parameters of thedriver chip so as to allow a large die including the BJT structure toturn on and/or off to compensate for the difference in for examplecarrier lifetime and/or doping levels.

The driver chip and other circuit component including base inductors andstorage capacitors may be mounted directly on top of a wafer comprisingthe BJT structure.

According to a further aspect of the present invention, there isprovided a method of manufacturing a bipolar junction transistor (BJT)structure, the method comprising: forming a base region of a firstconductivity type, wherein said base region constitutes a drift regionof said structure; forming first and second collector/emitter (CE)regions, each of a second conductivity type adjacent opposite ends ofsaid base region, wherein said base region is lightly doped relative tosaid collector/emitter regions; and forming a base connection to saidbase region, wherein said base connection is within or adjacent to saidfirst collector/emitter region.

The method may further comprise: etching the first collector/emitterregion; and forming a diffusion region in the etched region. The methodmay further comprise filling polysilicon in a trench to form the firstcollector/emitter region and/or to form a thin interfacial oxide region.

The method may further comprise applying an anisotropic wet chemicaletching of the first collector/emitter region with artwork aligned ateither zero degrees or 45 degrees to form a simultaneous undercut of anoxide and a self-terminating V-groove etch of contact holes.

The method may further comprise applying the anisotropic wet etching toform a bevel etch to control the edges of the BJT structure. The methodmay further comprise applying an electric field grading technique toreduce minority carrier injection from the collector/emitter regions.The method may further comprise forming a three dimensional or stackedstructure so as to give higher power ability and/or higher sensitivityand lower conduction losses.

The method may further comprise forming a recessed BASE contact so thatthe electrodes on collector/emitter regions can form the threedimensional or stacked structure.

According to a further aspect of the invention, there is provided anactive rectifier comprising:

-   -   a power bipolar junction transistor (BJT), having a first and        second input/output (I/O) connections and a base connection;    -   first and second rectifier terminals, wherein said first I/O        connection of said BJT is coupled to said first rectifier        terminal, wherein said second I/O connection of said BJT is        coupled to said second rectifier terminal;    -   a driver oscillator to provide a two phase drive waveform having        a first (on) portion and a second (off) portion;    -   at least one controllable switch controlled by said driver        oscillator and coupled between said second rectifier terminal,        said base connection of said BJT and said second I/O connection        of said BJT, to selectively route current from said second        rectifier terminal between said second I/O connection of said        BJT and said base connection of said BJT;    -   wherein said driver oscillator controls said controllable switch        to route said current from said second rectifier terminal        between said base and second I/O connections of said BJT in        proportion of a ratio of durations of said first and second        portions of said drive waveform.

The second I/O connection of said BJT may be coupled to said secondrectifier terminal via a filter, and wherein said filter may comprise acapacitor such that a connection between said second I/O connection ofsaid BJT and said second rectifier terminal is via said capacitor.

The active rectifier may further comprise an inductance between saidsecond rectifier terminal and said base connection said BJT to storecurrent for said base connection whilst said controllable switch isrouting current from said second rectifier terminal away from said baseconnection of said BJT.

The controllable switch may comprise a first controllable switch coupledbetween said second rectifier terminal and said second I/O connection ofsaid BJT and a second controllable switch coupled between said secondrectifier terminal and said base connection of said BJT and; and whereinthe two phase drive waveform may comprises first and second waveforms,said first waveform having an on portion corresponding to said firstportion of said two phase drive waveform, said second waveform having anoff portion corresponding to said second portion of said two phase drivewaveform, wherein said first waveform controls said first controllableswitch and said second waveform controls said second controllableswitch.

The controllable switch may comprise a first controllable switch coupledbetween said second rectifier terminal and said second I/O connection ofsaid BJT and a second controllable switch coupled between said secondrectifier terminal and said base connection of said BJT and; and whereinthe two phase drive waveform may comprise first and second waveforms,said first waveform having an off portion corresponding to said secondportion of said two phase drive waveform, said second waveform having anon portion corresponding to said first portion of said two phase drivewaveform, wherein said first waveform controls said first controllableswitch and said second waveform controls said second controllableswitch.

The active rectifier may further comprise a boost converter to boost avoltage drop across one or more circuit elements coupled between saidrectifier terminals to provide a power supply for said drive oscillator.

The boost converter may be coupled across one or more circuit elementscoupled in an emitter circuit of said BJT

The active rectifier may further comprise an inductance between saidsecond rectifier terminal and said base connection said BJT to storecurrent for said base connection whilst said controllable switch isrouting current from said second rectifier terminal away from said baseconnection of said BJT; and wherein said boost converter may comprisesaid inductance, to boost said voltage drop, and said driver oscillatorsuch that said driver oscillator, and inductance together with said atleast one controllable switch form a boost converter to power saiddriver oscillator.

The active rectifier may be configured to use leakage current throughsaid BJT, or a high voltage current source device, or a resistor, toprovide power to bootstrap said driver oscillator of said boosterconverter.

The on portions of said first and second waveforms are non-overlappingsuch that there is a dead time between said on portions; the activerectifier may further comprise a power harvesting device or Schottkydiode coupled to a connection between said second rectifier terminal andsaid second I/O terminal of said BJT to harvest power from said voltagedrop during said dead time.

The first I/O connection of the BJT may be a collector connection andthe second I/O connection of said BJT may be an emitter connection.

The ratio of durations of the first portion to the second portion of thetwo phase drive waveform may be less than 1:1.

BRIEF DESCRIPTION OF THE EMBODIMENTS

These and other aspects of the invention will now be further described,by way of example only, with reference to the accompanying figures inwhich:

FIG. 1 illustrates an example of a vertical cross-sectional structure ofa double-gated device in which (A) illustrates a dual-base version, (B)illustrates a single-base version and (C) illustrates an alternativesingle-base version;

FIG. 2 a illustrates a drive circuit used for the devices of FIG. 1;

FIG. 2 b illustrates a drive circuit used for the bi-directional BJTdevices of FIG. 1;

FIG. 2 c illustrates a power scavenging circuit;

FIG. 3 a illustrates hole current densities when operating as per FIGS.1 and 2;

FIG. 3 b illustrates electron current densities when operating as perFIGS. 1 and 2;

FIG. 4 a illustrates cross sections of an alternative BJT structure;

FIG. 4 b illustrates a polysilicon emitter system of an alternative BJTstructure;

FIG. 4 c illustrates an alternative BJT structure having a dual CE;

FIG. 5 is a schematic symbol of a BJT device in which (a) illustrates aP-channel JFET being effectively in series with the base terminal, (b)illustrates non-encroachment of donor atoms in channel and (c)illustrates encroachment of donors in channel;

FIG. 6 illustrates a driver circuit;

FIG. 7 illustrates a concept view of an alternative transistorcomprising multiple parallel connected stripes and metallisationtogether with field-plate extensions for increased breakdown voltage;

FIG. 8 illustrates an array of chips in which the chips are inter-wiredusing a flex-pcb and wire-bonded to the individual die;

FIG. 9 illustrates a layout of a 3D stacking of devices (folding) withfacility to increase surface area when even higher currents arerequired;

FIG. 10( a) shows an alternative route to a definite PNP input stage forthe BASE compared to the arrangement shown in FIG. 5;

FIG. 10( b) shows an alternative PNP input stage in which devices arestacked back-to-back.

FIG. 11 is a 3D view of a minimal unit stack which can be scaled in X, Yand Z;

FIG. 12 illustrates structures which enable an either/or choice ofwireless/wired and has an additional advantage of furnishing power tothe attached device in wired-mode without having to break the wires whena node is attached to the network;

FIG. 13(A) is a schematic bootstrap/boost voltage circuit diagram whichshows a +Ve voltage conduction of the AC transistor;

FIG. 13(B) is a schematic bootstrap/boost voltage circuit diagramshowing the two time portions of the PWM cycle (1) and (2) feedingenergy from VCE1 to a total-loss circuit;

FIG. 13(C) is a schematic bootstrap/boost voltage circuit diagram whichis operational when I_LOAD is negative;

FIG. 14 illustrates process steps of a bi-directional BJT device(JFET-base transistor) using Nitride;

FIG. 15 illustrates process steps of a bi-directional BJT device(JFET-base transistor) using oxide only;

FIG. 16 illustrates process steps of a bi-directional BJT device(BJT-base transistor) using oxide only;

FIG. 17 illustrates the processing steps of manufacturing thebi-directional device (BJT-base transistor) using a single mask in {100}and {110} etching methods;

FIG. 18 illustrates an alternative single mask scheme with self-limitingcontact depth;

FIG. 19 illustrates singulation/bevel/passivation steps for thebi-directional BJT device;

FIG. 20 illustrates electric field distributions in a bi-directionaldevice;

FIG. 21 illustrates the doping concentrations in a bi-directional BJTdevice;

FIG. 22 illustrates an array of CE1 stripes;

FIG. 23 illustrates a solid state relay module including a ‘slab’ typeinductor for bootstrap DC-DC;

FIG. 24 (A) is an illustration of a bi-directional BJT device (BJT PNPbase) in an off-state in which A) illustrates zero volts in terminals;B) illustrates CE2 with a positive voltage and other terminals at zerovolt, and C) illustrates CE2 with a negative voltage and other twoterminals still at zero volt.

FIG. 24 (B) is an illustration of a bi-directional BJT device in anon-state in which D) illustrates CE2 at +0.1V, CE1 at 0V and BASE at+0.6V; E) illustrates CE2 at −0.1V.

FIG. 25 (A) illustrates the off-state operations of an alternativebi-directional BJT device in which A) illustrates all terminals in zerovoltage; B) illustrates CE2 with a positive voltage and other terminalsat zero volt, and C) illustrates CE2 with a negative voltage and othertwo terminals still at zero volt;

FIG. 25 (B) an illustration of an alternative bi-directional BJT devicein an on-state in which D) illustrates CE2 at +0.1V, CE1 at 0V and BASEat +0.6V; E) illustrates CE2 at −0.1V.

FIG. 26 illustrates an arrangement of switches which operate in twophases in which A) illustrates positive inductor charging phase, B)illustrates positive inductor discharging phase, C) illustrates negativeinductor charging phase, D) illustrates negative inductor dischargingphase, E) illustrates initial bootstrap circuit, F) illustrates chargepump circuit, G) illustrates base drive circuit;

FIG. 27 illustrates a driver circuit and the associated voltage andcurrent waveforms in which A) illustrates multi-output inductive basedrive, B) illustrates example voltage waveforms, C) illustrates basepulse using pre-charge and discharge, (D) illustrates base fingerdriver, (E) illustrates alternative base finger driver, (F) illustratesinductor driven base waveform, (G) illustrates multiphase operationsingle-base connection, (H) illustrates base on/off controlled by PWM 1.

FIG. 28 illustrates a transfer curve of current vs. PWM value (0-255range) for one path showing that discontinuous current drive is highlynon-linear;

FIG. 29 is a schematic diagram of a digital current mode driver;

FIG. 30 illustrates cross-sections of a semiconductor device structuresin which A) illustrates a cross section and equivalent circuit of astandard IGBT, B) illustrates a cross section and equivalent circuit ofan alternative BJT, C) illustrates a cross section of an alternativeBJT, (D) illustrates a cross section and equivalent circuit of analternative IGBT, (E) illustrates the doping profile of the device ofFIG. 30 (D), (F) illustrates Beta vs Current Density waveform, (G)illustrates a top view and bottom view of a full die according to thedevices above;

FIG. 31 illustrates schematic circuit diagrams in which A) illustratesmain current path with a positive voltage, B) illustrates main currentpath with a negative voltage;

FIG. 32 shows a matrix converter system topology for low cost and highreliability using the transistor structures and driver techniquesdescribed in the previous embodiments in which A) illustrates variablefrequency matrix-converter drive system, and B) illustrates tripledriver module.

FIG. 33 (A) illustrates an example of a driver chip mounted to a powertransistor using an interposer flex-PCB;

FIG. 33 (B) illustrates an example of a programmable PWM skew circuit;

FIG. 34 illustrates an example of a low leakage relay switch;

FIG. 35 (A) illustrates an alternative scheme to a standard CMOS processto optimise it for the role of driver especially of NPN versions of thepower transistor where most of the PWM conduction current is via NFETdevices to/from 0V.

FIG. 35 (B) illustrates a simplified synchronous rectifier system forisolated power and data to/from driver IC using the CMOS chip;

FIG. 36 illustrates an example of a 3-phase inverter using DC bus andsynchronous mains rectification;

FIG. 37 (A) illustrates an active diode concept;

FIG. 37 (B) illustrates an example of 10 amp forward conduction(ignoring inductor and C1 ripple current);

FIG. 37 (C) illustrates an integrated version of a vertical B2 device;

FIG. 37 (D) illustrates a hand-made demonstration of self-resonantcircuit;

FIG. 37 (E) illustrates an air-core or ferrite inductors;

FIG. 37 (F) illustrates a Veroboard construction;

FIG. 38 (A) illustrates a C2 device circuit;

FIG. 38 (B) illustrates CMOS integration of an I2 device resulting inC2;

FIG. 38 (C) illustrates drive waveforms;

FIG. 38 (D) illustrates a plan view of a standard cell of C2 device;

FIG. 39 (A) illustrates D2-control CMOS basis arrangement;

FIG. 39 (B) illustrates D2-control for T2 transistor;

FIG. 39 (C) illustrates B2 or T2 die having 3.3 mm×3.3 mm dimensions;

FIG. 39 (D) illustrates a stacked die on printed-conductor substrate;and

FIG. 39 (E) illustrates an encapsulated bridge rectifier design.

DETAILED DESCRIPTION OF THE EMBODIMENTS An Example of a Bi-DirectionalTransistor Design

What follows is a general non-limiting explanation of the concepts andan initial design which may not be subject to well-known optimisationtechniques for highest gain, highest voltage withstand ability.

An NPN type structure may be described but certainly a PNP structure isalso possible by reversing the doping systems. Where reference is madeto diffusion, ion implantation is also an option and so on.

FIG. 1(A) shows an example of the vertical cross-sectional structure ofa double-gated device. On each face of a lightly doped P− wafer arerelatively-deep diffused Nmed (doping ranging from 10¹⁸ cm⁻³ to 10²¹cm⁻³) regions to form the CE (Collector-Emitter) regions 100. These Nmedregions are routed to metal contacts in the usual way via a highly dopedN+ diffusion to make the CE electrodes 105. Metal contact to the P−wafer is achieved with a relatively shallow P+ diffusion to make a Baseelectrode 110, 115.

Useful for working operation of the “Invisible Base” concept of JFETsecond transistor is the management of several design factors within acontrolled range. When the parameters are within a solution-space thenthe Nmed regions fully deplete the P− silicon around the Base contactarea and stop what would otherwise be a direct conduction path from theupper Base1 electrode 110 to the lower Base2 electrode 115.

The Base may be “invisible” because of this depletion but stillfunctions like a BJT Base when forward biased. The transistor may alsowork even when the P− base region is impinged by the N doping to adegree. In this case the operation of the base ‘channel’ may besomewhere between the physics of PNP and a Pch JFET.

Below are listed variables are given for an initial working solution fora 1200V device with a 50 mV on-voltage and a current gain of 15. Defaultdoping concentration profiles are assumed.

-   -   Silicon wafer thickness (120 micron)    -   dist_x_total_microns=50    -   dist_x_spacing_microns=1.0    -   dist_x_base_diffusion_microns=2.5    -   dist_y_shallow_junction_microns=1.0    -   dist_y_deep_junction_microns=5.0    -   substrate doping=1.5e14 cm−3    -   Nless doping=1e18 cm⁻³    -   N+ doping=1e19 cm⁻³    -   P+ doping=1e20 cm⁻³

The pitch of repeating patterns would be at “dist_x_total_microns/2”intervals and metallisation would connect the multiple stripes togetherin the normal way to make a larger device. The width in the TOADsimulations was 10,000 microns.

FIG. 1 also shows the electrical symbol for the structures.

The dual-base structure of FIG. 1(A) is intended to be driven bytransformer-coupled base windings. Minority carriers are injected on thetop and bottom of the structure through the P/N, i.e. Base/CE forwardbiased junction increasing conductivity between CE1 and CE2 terminals.The symmetry gives equal current gain (hFE) in both the forward andreverse polarity power conduction directions.

A single base structure of FIG. 1(B) when driven according to FIG. 2 a(but with Base2 terminal omitted) injects carriers only on the top sideand so hFe is approximately halved when not switching in the preferredCE voltage polarity quadrant. However, this structure has the bigadvantage of a single base drive circuit (directly coupled) and does notrequire masking and patterning on both sides of the wafer as is the casewith the alternative designs.

Finally, in one embodiment, the structure of FIG. 1 (C) is similar buthas two CE electrodes 130, 135 displaced laterally on the lower side ofthe silicon and relies on minority carriers being injected from ‘above’or the top side by the dedicated base and emitter region on the topside. This design may suit lower voltages and thinner silicon wafers.

In all cases above it will be appreciated that, in the bi-directionaltransistor design, the Base (i.e. the P of the NPN structure) is workingas the Drift region and supports the full ‘off’ voltage rather than theusual configuration of an NPN transistor where the collector (N regionof an NPN structure) would act as the voltage-supporting drift region.

All the structures described above have much lower losses and hencehigher efficiency than the standard IGBT device.

FIG. 3 shows the Hole and Electron current densities when operating asper FIGS. 1(A) and 2 a.

FIG. 2 b illustrates a drive circuit 200 used for the bi-directional BJTdevices of FIG. 1. The drive circuit 200 has a microprocessor control ofthe base current and can give complete safe-operating area, shortcircuit protection, zero-crossing on and off all defined by software. ASTM32F373 microcontroller for example is able to control up to 6switches at an added cost of $0.50 per switch. It has flash ROM for datalogging and UART I/O for communication.

In FIG. 2 b, MOSFETs Q2 and Q3 drive a small inductor L using PWM outputand synchronous rectified buck-converter technique to efficiently createthe low (0.7V typical) base voltage and current to switch the maintransistor Q1 (or the bi-directional BJT) on. Q4 is a quick-turn-offdevice for Q1 and can impose a negative base bias which helps toincrease the breakdown voltage of Q1. Rsense1 and the ADC channels ADC4,ADC3 give feedback to the control program as to the instantaneous basecurrent and base voltage. Rsense2 with ADC2 measures the emitter current(which includes base current which can be subtracted out digitally).ADC1 via the protection resistor Rprotect measures the transistorvoltage drop (VCE1−CE2) when switched on. An algorithm can adjust thePWM ratio until the targeted voltage drop (VCE1−CE2) is maintained. Justsufficient base current will be used which can prevent deep saturationof the main transistor which otherwise renders it slow to turn-off.

ADC0 with voltage divider formed by Rdiv1, Rdiv2 allows thezero-crossing time of the mains waveform to be detected (for optionalzero-crossing synchronised on/off of the power switch) and for thesmart-appliance metering application the total power delivered to theload is given by multiplying this value by the through collector/emittercurrent determined previously.

Description of Power Scavenging Concept (Illustrated in FIG. 2 c)

A detailed circuit is described later but in principle thebi-directional BJT device can get its base current from the voltage dropacross itself while it is switched on providing that the Hfe (currentgain) is sufficiently high.

For example with 10 A passing through the switch, an assumed Hfe(current gain) of 20, and a VCE1−VCE2 drop of 0.15 volts and the Vbe toswitch the device on is 0.7 volts.

The power used in the base is 10 A/20*0.7 volts=0.35 W, while the powerloss over the switch is 10 A*0.15V=1.5 W. Extracting 0.35 W with anefficient DC-DC boost converter operating at the 0.15V requires ahigh-voltage switch Q11 which protects the low-voltage circuits when themain transistor is off. For example, 2.333 amps is needed from this0.15V source to power the base. If Q11 itself is of BJT typeconstruction then it too needs a source of a lower base current from thescavenging power supply.

To make things easier and in pursuit of the lowest overall losses theintelligence of the microprocessor can be used so that Q1's VCE1−VCE2drop is deliberately higher while switched on during the low-currentportions of the mains cycle in order to extract and store energy intothe Vdd, Vss capacitors ready for implementing lower voltage drop in Q1during the peak voltage times.

Description of the Charge Control Model:

With a microprocessor in control of the switching, and digital feedbackof all the analogue quantities in the circuit a charge-control model canbe executed in order to keep the power transistor switched on in themost efficient way without overdriving the base. Generally, a measure ofthe VCE1−VCE2 voltage drop is taken and if lower than a pre-set target,e.g. 0.1 volts, then more base current is commanded from the PWM.Knowing that internal charge is building up on the base/CE junctions andthe capacitance of these junctions and the required minority chargeneeded to support a particular switched current, the quantity andduration of this base current boost can be regulated in order tointercept the demand current through the device as it is seen to rise.Similar algorithm can be used for reducing currents. The algorithm canalso take into account the recombination lifetime to have a constantestimate of the charge available for conduction.

Flash Memory

With a self-writable flash memory in the microprocessor, each powertransistor can have permanently stored a calibration area setup aftermanufacture during test and referable when operational to improve theaccuracy of the algorithms and the reported measurements of the device.

Alternative Fabrication Technique

An example of the fabrication technique of the devices of FIGS. 1 and 2is described below.

Etched Trench:

From additional simulations it is found that an etched trench withvertical sidewall in the silicon, when diffused, gives a FET-‘base’structure of higher performance than a simple diffused planar junction.

It is also more economical in terms of time and equipment utilisation toetch a trench and make a shallow diffusion (˜30 minutes total) than adeep thermal diffusion which can take many hours. It also allows asharper edge to the diffusion profile to be achieved.

FIG. 4 a shows the structure and the identified parts.

Another option is a “Polysilicon Emitter” system shown in FIG. 4 b inwhich a heavily doped N+ polysilicon is used to fill the trench formingthe emitter. Similar N+ polysilicon emitter formed on the bottom sidealso. Polysilicon emitters have higher gain at higher current densitiesdue to the barrier for hole injection (from the base into the emitter)formed by an inherent oxide layer which forms between the silicon andpolysilicon.

Equivalent Circuits of the IBT Device (or the Bi-Directional BJTDevice).

FIG. 5 is a schematic symbol of the bi-directional BJT device.Equivalent circuits are shown for the structures discussed previously.In the first case (in FIG. 5( a)), a P-channel JFET is effectively inseries with the base terminal. This happens when the N+ does not diffuseall the way into the channel region under the base contact. In thesecond case (in FIG. 5( b)), where some donor atoms do make it into thechannel, operation of the base drive does not immediately fail.Operation becomes that of a lightly-doped base PNP transistor in serieswith the main P− base region. Too much encroachment may reduce theoperating efficiency of the device and can slow down turn-off somewhatbut it remains operational (in FIG. 5( c)).

Solid-State Relay Replacement (with Reference to FIG. 6)

FIG. 6 illustrates a driver circuit 600. The circuit 600 can serve asdrop-in replacement of a standard solid-state relay. The circuit may beable to operate from just 2 power terminals, i.e. the contact terminals.The power available from the signal side of the relay (e.g. 5 mA@5volts) is insufficient to power the IBT base even through a transformer.

A power scavenging system is required in lieu of another source ofexternal power.

Referring to FIG. 6, when the IBT transistor is off, a high bleedresistance can tap a few micro-amps (which appears as slightly elevatedleakage current into the load) to power an ultra-low-powermicrocontroller circuit which can boot up and run at a low Khz-typefrequency.

A storage capacitor C on +VDC line has sufficient charge such than whenthe IBT is required to be turned on, there is enough energy to drive thebase at least initially.

With the IBT turned on there is now a low voltage (the voltage drop)across the contact terminals CE1 and CE2 in proportion to the currentconducted multiplied-by the ‘On’ resistance of the transistor. Thisvoltage-drop is a parasitic effect but it can be used to extract asource of power for maintaining base current in the IBT. There are twoimmediately apparent options for this. First there is a special dual-CE2structure as shown in FIG. 4 c. Secondly, an additional IBT operating atlower current density (and therefore low voltage-drop) can tap the CE2voltage and conduct it onto VTAP as an input to the DC-DC converter.

An example of option 1: Device terminal CE2Y will also have a voltage ofapproximately CE2X (since the P− region is filled with minorityconductive carriers). The voltage at CE2X can be boosted to give acontinuous DC power source for powering the base.

Depending on whether the voltage drop is +Ve or −Ve (detected by acomparator in the microprocessor) the microprocessor activates theswitches in the following logic sequence then repeats.

An example of option 2: If the main transistor has a gain of 25 atVCE1/CE2=0.1V at 5 A/cm² then it needs 0.2 amps/cm²@0.7V for the basewhich is 1.4 A/cm² from 0.1V. If equal area is given over to the tap IBTit will be operating at 3× lower current density and proportionallylower voltage drop and higher HFE (gain) than the main transistor butstill VTAP would likely be 70 mV instead of 100 mV. Iteratively solvingthis (and including the base current for the second IBT) it isanticipated that the IBTs can be self-powered with a 2× increase intotal device area and 1.5× higher voltage drop. The voltage drop isstill around 10× lower than comparable switching device on the market.

In both cases the circuit self-adjusts to the biasing conditions, forexample that the current through the switch rises, then so does thevoltage-drop over IBT1, this gives more voltage available to VTAP andmore power to the base drive which in turn helps to lower the VCE1/CE2saturation voltage.

+Ve voltage drop at CE2. Voltage needs to be boosted by ˜10×.

-   -   Phase1. Agate=0, Bgate=1, Cgate=0, Dgate=1. Duration=100 uS        example.    -   Phase2. Agate=1, Bgate=0, Cgate=0, Dgate=1. Duration=10 uS        example.    -   [repeat]        −Ve voltage drop at CE2. Voltage needs to be inverted and        boosted by 11×.    -   Phase1. Agate=0, Bgate=1, Cgate=0, Dgate=1. Duration=110 uS        example.    -   Phase2. Agate=0, Bgate=1, Cgate=1, Dgate=0. Duration=10 uS        example.    -   [repeat]

For option 2, the bases of the two IBT transistors could be driven byindependently controlled PWM/Inductor circuits instead of both basesbeing driven together. This facilitates turn on of the Q2 independentlywhich could be used in addition to the bleed circuit to gather largercurrents via the load resistance by switching on around thezero-crossing times of the mains waveform. Low voltages could beefficiently gathered in this way and for many loads (e.g. heaters, largemotors) the small additional ‘leakage’ would not affect them.

Low Cost Manufacturing

One possible starting material is P− high-minority-lifetimemono-crystalline solar wafers which can be readily sourced in a range of50 u to 300 u thickness. The thickness and doping depend on chosenwithstand voltage of the device (thicker, lower doped for highervoltage). At current prices, processed P− silicon wafer panels for solarpanels having been through the following processing steps: Etch,Diffusion P and N, Contacting and Passivation retail for $0.02 per cm².Operating at 2.5 A/cm² for very low voltage drop and high gain, 10 cm²of silicon (20 cm² for self-powered) might be used for a 25 A device.The high-volume target silicon cost could be as low as $0.50 c for a 25A 1000V 0.1 W per amp loss SSR using these techniques.

A concept view of an IBT transistor comprising multiple parallelconnected stripes and metallisation together with field-plate extensionsfor increased breakdown voltage is illustrated in FIG. 7

In order to accommodate the large silicon areas required for lowestloss, a 3D stacking technique would be used. Since each slice is only0.2 mm thick, a 25 A SSR could be fitted into an area of 20 mm×10 mm×2mm. No special additional heatsink (˜$2 cost) would be required.

FIG. 8 illustrates a chip in which the chips are inter-wired using aflex-pcb and wirebonds to the individual die. FIG. 9 illustrates alayout of a 3D stacking of devices (folding) with facility to increasesurface area in case even higher currents are required (inter-wiringomitted for clarity).

The package may be coated in encapsulant for environmental protection.

Minority Lifetime Choices

High minority carrier lifetime is important in the P− region to get highHFE (gain).

However, shorter lifetime could be a target for the N+ regions wherehigh doping also tends to kills HFE at low current levels—this help toincrease breakdown voltage.

Alternative Transistor Construction

FIG. 10 shows an alternative route to a definite PNP input stage for theBASE (re. FIG. 5). This scheme can have lower charge-removal during turnoff relative to JFET, but has some advantages in high-current density,and can allow a simpler 3D stacking system to increase current rating ina given footprint area. A typical process is given for an R&D lab. Theillustration is a minimal unit drawn as a half (right hand half)cross-section of a cylindrical symmetric design.

-   -   Firstly a P− doped wafer has 20 u deep Phosphorous diffusion on        both sides. This gives N+ doping levels near the surface and N−        towards the bottom of the diffusion. A polysilicon layer or        silicon-carbide hetero-junction layer can be added on both sides        to create polysilicon emitters for higher gain if desired.    -   Then an oxide layer is grown on the top side of the wafer (or an        insulator layer is deposited)    -   Openings in the insulator layer are made using        photoresist-expose-etch.    -   A silicon-etch (e.g. KOH) is performed to make a trench for the        BASE electrode in the places defined by the previous openings.        The insulator acts as a mask. Undercutting is helpful.    -   A boron diffusion is performed. The insulator acts as a        diffusion blocking mask meaning that only a thin shell of P+        diffusion is made at the walls of the trench.    -   Now a second set of openings are made in the oxide in the        regions where contact to the N+ is to be made.    -   Aluminium is sputtered from above. This gives        mutually-unconnected metallisation connection to the BASE and        CE1 regions. Lower cost alternatives such as metallic pastes        could be used.

For higher current such devices can be stacked back-to-back as shown inFIG. 10 b. Inserting metal sheet between each layer in the stack canobtain electrical connection. The stack could be pressed and fired tomake thermo-compression bonding.

FIG. 11 is a 3D view of a minimal unit stack which can be scaled in X, Yand Z. A BASE connection will contact to two silicon die using solder,conductive paste or thermo/compression bonding. Similarly the CE1 andCE2 copper sheets will each connect to two different die using perhapssintered silver powder or high temperature solder—on the CE1 sideprobably screen printed onto the copper plates in a pattern to match theCE1 openings.

The final connection scheme to the outside world is determined by theshape of the copper sheets which are stamped and bent to make a kind oflead-frame.

A thermo-compression process can ‘sinter’ the stack together attemperatures between 250° C. and 400° C. typically, or conventionalhigh-lead solder can be used.

A final encapsulation with plastic (not shown) will allow a SMDelectrical mounting of the device onto a PCB.

Alternative Signalling Scheme to Communicate/Power the SSR Modules

Bluetooth and Bluetooth Low Energy are examples of communication systemfor short-range RF data exchange. This system could be used as-is tonetwork multiple solid-state-relays (SSRs) of the type (describedpreviously in this document) by using for example a Bluetooth-capableSOC including Microcontroller from, for example, Nordic SemiconductorInc or Texas Instruments as an upgrade for the system microcontrollermentioned previously. Normally an antenna at each end of such acommunications link permits wireless connectivity. These antennas can berealised from a specific PCB trace pattern whose layout is specific andtuned to the radio frequency involved—typically 2.4 GHz. For industrialcontrol networks wireless is not often used due to the possibility ofinterference and security issues. A wired network is used instead.

FIG. 12 illustrates structures which enable an either/or choice ofwireless/wired and has an additional advantage of furnishing power tothe attached device in wired-mode without having to break the wires whena node is attached to the network.

A twisted pair electrical wire 1210 (e.g. UTP or STP) is capable oftransmitting 2.4 GHz RF signals for lengths of 15 meters with acceptableattenuation. Underneath this RF frequency can be a low-frequency powerwaveform e.g. ˜20 KHz created by a central generator.

On an intelligent SSR node which is to tap this power and the RFcommunications, the UTP cable is locally ‘untwisted’ to produce a largerthan usual loop. This loop is placed inside a hinged magnetictransformer 1215 which is a type of planar transformer for the LF powerfrequency and forms the primary. The hinge 1220 is then closed tocomplete the magnetic circuit. The secondary side of the transformer isa PCB trace on the SSR module and has terminals labelled X1 and X2.Rectifiers and filter capacitors on the secondary permit extraction ofpower from power waveform.

This configuration allows for solder-free insertion and removal itemsinto the network and gives 100% electrically isolation. The ferrite usedfor the transformer needs to have high permeability to the LF and act asa transmission-line transformer for the RF.

If designed correctly, the PCB trace for the secondary of thetransformer is also able to couple the typically 2.4 GHz RF signal fromthe UTP cable loop and into a Bluetooth RF chip on the SSR pcb 1225.

An inductor-capacitor network and rectifiers can separate the LF powerfrom the RF frequencies.

Signals X1 and X2 from the pcb-transformer become VDD, VSS and RF1, RF2.

With sufficient screening in the RF components and transformer, anetwork of many nodes would only be responsive to signals in the UTPcabling and not very sensitive to general external Bluetooth signals inthe building. This allows high reliability signalling and no possibilityof eavesdropping. It also means that bandwidth increases when runningmultiple independent RF-UTP networks which run without collisions.

At the end of the UTP network cable a terminator 1230 is added. Thisgives an LF return current path which effectively puts all the Nodes inseries to the AC power generator. Higher number of nodes requiresproportionally higher voltage LF power signal.

With the hinged PCB transformer open, the unit would operate in a normalBluetooth wireless mode which is good for low-security networks andduring development. In this mode power would have to come from aself-powered scheme as described previously.

Simulation of Bootstrap/Boost Circuit

FIG. 13 is a schematic bootstrap/boost voltage circuit diagram in whichthe current and voltages seen by the bootstrap/boost voltage circuitoperating on CE1 terminal of the AC transistor is illustrated. Thecircuit is based on that described in FIG. 6.

FIG. 13( a) shows a +Ve voltage conduction of the AC transistor. VCE1node has a capacitor C1 which is charged up by the I_LOAD currentpassing through the transistor. VCE1 works like a low voltage powersource. All the power is extracted through the inductor and passed ontothe BASE to replenish recombination and other losses in the base regionof the AC transistor.

FIG. 13( b) simplifies the circuit to show the two time portions of thePWM cycle (1) and (2) feeding energy from VCE1 to a total-loss circuit(which in practice is replaced by the power supply load the controlsystem—including microcontroller and the BASE drive). The arrangement ofFIG. 13( b) is for operational when I_LOAD is positive. FIG. 13( c) isoperational when I_LOAD is negative which generally requires the voltageinversion necessary.

Closed loop regulation of the bootstrap/boost mode can be implemented bya digital algorithm operating with input ADC results from all therelevant voltages and current monitor points (of the kind shown inprevious diagrams) and outputting to appropriate PWM control of basedrive (similar to FIG. 2 b).

Appendix 1 lists a python-code simulation which works with both +Ve and−Ve loads—determined as per FIG. 6 (VTAP is same as VCE1 of FIG. 13).The algorithm first locks up the loop to achieve a current balance. WhenI_LOAD matches the average current in I_INDUCTOR then VCE1 voltage isstable (does not rise or fall).

An additional goal for the loop is regulation of the VCE1 to a targetvoltage. For example +0.1V or −0.1V is a good target. Given thisvoltage, the algorithm-locked DC-DC conversion process will extractpower to provide to the BASE a current of approximately 1 Watt per10-amps of I_LOAD current. Relating this to a typical VBASE of 0.75Vgives a base current 1.333 A and corresponds to a “forced-beta” of about7.5. The system automatically gives a base current in proportion to theload current without algorithm intervention but the VCE target voltagecan be algorithmically changed dynamically if the transistor is seen(through ADC using circuits explained previously) coming out ofsaturation.

Digital Fuse/Circuit Breaker

The current-limiting and self-bootstrap power allow the completeintelligent device to operate as a two-terminal device. The unit caninitially scavenge deliberate leakage current and periodically wake-upthe microcontroller as has been described previously. When switching to“On” conduction mode it can run from the boost/bootstrap power andmaintain very low total losses compared to a standard fuse. The boost involtage is sufficient to power normal control circuits even though themeasured voltage across the two-terminal ‘Fuse’ would be 100 mV or less.

Such a fully programmable digital fuse can stand in for a traditionalfuse in any application and has superior ‘clearing’ speed if needed orcan be programmed to emulate any type of slow, medium or fast fuse typewith a programmable ‘trip’ point. Obviously it has the advantage over astandard fuse in that it is not destroyed when activated and couldself-reset with time, or power cycling events etc.

APPENDIX 1 Source code for DCDC +/− converter simulator #!/usr/bin/envpython # # - this one tries to close the loop # - can do +VE or −VE #with INVERTED_MODE # # In inverted mode simulates the proper mosfetsequences. # All currents are the right sign. # INVERTED_MODE : # Onlything that gets done is to VBOOST which remains positive # and to copewith that, the switches are simulated. # the opposite sign of VBOOST isused in all the calculations # relating to how it affects the change incurrents. Plus the current into C2 gets swapped. L=10e−6 IL=0.0 VCE1_NOM= 0.1 VBOOST_NOM = 3.3 VCE1 = 0.0 ;#- voltage drop to work off VBOOST =3.3 ;#- running voltage on C2 VZENER = VBOOST RZENER = 0.5 fpwm =30000.0 tcycle = 1.0 / fpwm ;#- cycle time time=0 #-work out initial pwmratio for no net current accumulation PWM_ratio_NOM = 1.0 −(VCE1_NOM/VBOOST_NOM) ;#- doesnt need the +1 because current calculatedlater as ramp down is from difference between VCE and VBOOST print“PWM_ratio_NOM=”,PWM_ratio_NOM Iaverage_VBOOST=0 IL_peak_max=0IL_peak_min=0 C1 = 1000e−6 ;#- VCE1 capacitor C2 = 100e−6 I_CE1 =10.0;#- amps coming from the switch I_BASE=0.0 ;#- actually the currentrunning around the zener in this case#---------------------------------------------- defrun_for_secs(seconds) : globaltime,IL,Iaverage_VBOOST,IL_peak_min,IL_peak_max,PWM_ratio,tcycle,VCE1,VBOOST,INVERTED_MODE tend = time + seconds while True : print “” ton = tcycle *PWM_ratio toff = tcycle − ton #Ramp up in current IL += ton * VCE1 / L;#- current increased during the on time IL_peak_max = IL #- work out theaverage current during the buildup phase Iaverage_VBUILDUP =((IL_peak_max + IL_peak_min) / 2.0) * (ton / tcycle) ;#- with ton=0,this will be zero VCE1 += (I_CE1 − Iaverage_VBUILDUP) * tcycle / C1 ;#-recalc capacitor voltage #if VCE1 <0 : VCE1=0.0 #Ramp down in current ifINVERTED_MODE == True : #invert mode, ramp down rate given only byVBOOST (other side switched to gnd) IL −= toff * −VBOOST / L else :#normal mode, VCE1 reduces the I ramp down speed because it subtractsfrom VBOOST IL −= toff * (VBOOST − VCE1) / L #NOTE, IL is allowed to gonegative ---- using a switch, not a diode - it can pump current bothways IL_peak_min = IL Iaverage_VBOOST = ((IL_peak_max + IL_peak_min) /2.0) * (toff / tcycle) ;#- average current through the boot (diode)portion print PWM_ratio print“uS=”,int(time/1e−6),“VCE1=”,VCE1,“VBOOST=”,VBOOST, “toffnS=”,int(toff/1e−9) print“~~IL_peak_max=”,IL_peak_max,“IL_peak_min=”,IL_peak_min,“I_CE1”,I_CE1,“Iaverage_VBOOST=”,Iaverage_VBOOST,“Iaverage_VBUILDUP”,Iaverage_(—)VBUILDUP if VBOOST > VZENER : I_zener_loss_on_C2 = (VBOOST − VZENER) /RZENER else : I_zener_loss_on_C2=0.0 ;#- no current if havent passedzener threshold if INVERTED_MODE == True : VBOOST −= Iaverage_VBOOST *tcycle / C2 ;#- simulate the switches and swap ba else : #normal VBOOST+= Iaverage_VBOOST * tcycle / C2 #add the resistive drop VBOOST −=I_zener_loss_on_C2 * tcycle / C2; #in inverted mode, I doesnt continuefrom C1 when boosting, comes from GND instead if INVERTED_MODE == False: #Only for normal mode is VCE1 affected during the boost time VCE1 −=Iaverage_VBOOST * tcycle / C1 ;#- the boost current also comes out of C1time += tcycle ;#- move time on if time >= tend : break#------------------- #gets the PWM which would bring about a deltacurrent change - based on current conditions of C1 and C2 voltages (VCE1and VBOOST) # - positive delta current is an increase current for theinductor # def calculate_pwm_for_a_delta_I (delta, scaling) : #see .mcdglobal L,tcycle,VBOOST,VCE1,INVERTED_MODE if VBOOST == 0.0 : VBOOST =1e−6 ;#- avoid /0 #See .mcd for slightly different formula ifINVERTED_MODE == True : PWM_ratio_for_no_I_change = ( (0 * L) + (tcycle * −VBOOST) ) / ( tcycle * (−VBOOST + VCE1) ) ;#-so can scale +/−around the zero point PWM_ratio_cancelling = ( (delta * L) + ( tcycle *−VBOOST) ) / ( tcycle * (− VBOOST + VCE1) ) ;#- which would be usable ifno scaling else : PWM_ratio_for_no_I_change = ( (0 * L) + ( tcycle *VBOOST) − (tcycle * VCE1) ) / ( tcycle * VBOOST ) ;#-so can scale +/−around the zero point PWM_ratio_cancelling = ( (delta * L) + ( tcycle *VBOOST) − (tcycle * VCE1) ) / ( tcycle * VBOOST ) ;#- which would beusable if no scaling relative_PWM_from_nochange_position =PWM_ratio_cancelling − PWM_ratio_for_no_I_change ;#- normalisescaled_relative_PWM_from_nochange_position =relative_PWM_from_nochange_position * scaling ;#- scale returnval =scaled_relative_PWM_from_nochange_position + PWM_ratio_for_no_I_change;#- then put back again #clamp if returnval > 1.0 : returnval = 0.999999if returnval < 0.0 : returnval = 0.000001 return returnval#------------------- #Iterate with various currents ~~~~~ #SET THETARGET VOLTAGE on VCE1 VCE1_target = 0.15 ;#- can be + or −VCE1_tune_resistance_ohm = 0.2 ;#- effective if VCE1_target < 0.0 :INVERTED_MODE = True #VCE1_target = 0.0 − VCE1_target ;#- Put it back tobeing a positive one else : INVERTED_MODE = False PWM_ratio =PWM_ratio_NOM old_VCE1 = VCE1 #Keep I_CE1 as positive number, will useINVERTED_MODE (when VCE1 is negative) to send currents the other way forI_CE1 in [10] : if INVERTED_MODE == True : I_CE1 = 0.0 − I_CE1 for cyclein range (450) : #Can work out what the I imbalance in C1 is by rate ofrise of voltage on it delta_I = C1 * (VCE1 − old_VCE1) / tcycle ;#- Ifnew voltage greater than old voltage then cap is risign from not enoughjuice taken out #So work out a duty cycle which will send towards zero#- see .mcd delta_I_gain = 0.75 #Think this an integrating effect.Result gives the delta to change the current by in each cycle based onthe voltage error VERROR = VCE1 − VCE1_target ICHANGE = VERROR /VCE1_tune_resistance_ohm ;#- if VCE1 is too high, this gets positiveprint “ICHANGE”,ICHANGE PWM_ratio_cancelling =calculate_pwm_for_a_delta_I (delta_I + ICHANGE, delta_I_gain ) print“cycle#”,cycle print “delta_I”, delta_I,“PWM_ratio_cancelling”,PWM_ratio_cancelling PWM_ratio =PWM_ratio_cancelling old_VCE1 = VCE1 run_for_secs(tcycle) print“------------------------------------------------------------------------------------------------------------------” print “- RESULT @ I_CE1 = ”,I_CE1 print “ VCE1 = ”, VCE1print“------------------------------------------------------------------------------------------------------------------”

Detailed Fabrication Techniques

The diagrams and notes are example methods of manufacturing thebi-directional BJT device.

The processes will mostly be with reference to a PNP input (main)transistor but the same techniques are applicable for the JFET version.

FIG. 14 illustrates the process steps of a bi-directional BJT device(with a JFET second transistor) using Nitride. These steps use acombination of Silicon Nitride and Silicon Dioxide as masking/insulatingmaterials. One advantage of these steps is that the lithography for twomasks can be done prior to the etching processes and the wafers do notneed to be returned to lithography part-way through production.

FIG. 15 illustrates the process steps of a bi-directional BJT device(with a JFET second transistor) using oxide only. As to the steps ofFIG. 14, these steps do not use Nitride but with the 2 oxidemasking/etch steps at different point in the manufacturing process.

FIG. 16 illustrates the process steps of a bi-directional BJT device(with a BJT second transistor) using oxide only. In these steps 2 oxidemasking/etch steps are needed

FIG. 17 illustrates the processing steps of manufacturing thebi-directional device (BJT-base transistor) using a single mask in {100}and {110} etching methods. In these steps, one oxide masking/etch stepis used. This is advantageous since there is no need for a second masklayer and it also provides alignment accuracy. The process relies on theanisotropic wet chemical etching of crystalline silicon using KOH, NaOH,TMAH, EDP or similar solutions. The contact holes for CE1 are smallopenings whose edges are controlled by the {111} planes to make 54degree inverted pyramids whose sidewalls can be calculated to come to apoint at a precise depth below the initial silicon surface. After this,etching stops which gives a controlled depth independent of furtherincreases of etch time. The trenches for BASE are oriented on the {100}direction and etch downwards at the same rate as laterally and givevertical sidewalls with 100% undercut ration. Etching is notself-limiting and is controlled using time.

FIG. 18 illustrates an alternative single mask scheme with self-limitingcontact depth. In this scheme unrestricted trench depth usingcrystallographic anisotropic etch is used. This has the 2D mask artworkfor forming a PNP-BASE transistor 1805 on a standard {100} orientedsilicon wafer. The wafer flat {110 plane} is aligned to the bottom edgeof the layout. Provided that the CE1 contact openings are small relativethan the desired trench depth then the CE1 contacts will only penetrateinto the very heavily doped top region of the N+CE1 diffusion. A 3Dresult from an etching simulator is shown for KOH as the etchant. Fromthe cutaway view the underside of the etched pyramid profile can beseen.

The trench depth is not limited for CE1 because the 2D outline forms aconvex shape bounded by {100} vertical planes (due to the 45 degreerotation of the artwork). This allows independent control of the trenchdepth determined by (etch-rate*time), and this BASE trench depth ischosen to cut into the phosphorous diffusion to a lower-doped regionwhere functionality of the PNP transistor is improved (see dopingprofile below). However, when Boron diffusion is applied to form theBASE of the composite device (i.e. the emitter of the PNP) it also dopesthe CE1 contact openings to ˜1e20/cm3 with boron atoms. This may notsufficient to over-compensate the previous Phosphorous diffusion of˜1e21/cm3 so the contact remains ohmic and only the nature of the N+ isseen in CE1 terminal. The next step would be to apply thermal Aluminiumevaporation where the oxide overhang (resulting from the undercut) makesindependent contact to the CE1 and BASE regions. If the gap between CE1stripes is minimised it might be possible for the aluminium to bridgeover the small gaps in the oxide to form a single CE1 terminal with noadditional wire-bonds.

FIG. 19 illustrates singulation/bevel/passivation steps for thebi-directional BJT device. To obtain a high breakdown voltage in asemiconductor material, it requires careful control of the edges of thedevice. At the edges and along the surfaces of the depletion regionleakage currents can arise which lead to much lower than expectedbreakdown voltage of the junctions. These issues have been solved inmany ways in the past but for AC devices there are complications in thatthe method is likely to be symmetrical. Reference“double_bevel_edge_termination_for_bidirectional_devices_(—)1974.pdf” isa paper from 1974 outlining the method of double-positive-bevel edgetermination where it was achieved using sandblasting. This same profilecan be achieved here with an anisotropic etch technique. First the waferis held by the front surface using a special vacuum chuck with rubbersealing strips. Then X and Y deep grooves are sawn with a diamond-coatedgrinding wheel from the rear—almost the entire way through the wafer.This is followed by anisotropic etching again from the rear side of thewafer to yield the required double-positive-bevel profile in about 20-50minutes of etching with 15% KOH at 100° C. Following the {111} planes upand down reveals a very smooth double bevel angle which is ultimatelyintersected and terminated. The bevel exists around whole outsideprofile of the die, including corners. To smooth the facets, a finalisotropic etch can be used. Etching stops at the SiO₂ layer. At the endof etching process, the devices are held together by just a thin SiO₂oxide layer and an aluminium layer which are on the front-side of thewafer and can be easily singulated—but first a passivation layer isneeded to cover the side profiles of the device. This passivation sealsout contaminants, controls the electrical charge on the layers andreduces surface recombination. For P-type silicon, use can be made of acommercial of alumina (Al2O3) passivation technique from solar waferproduction called atomic-layer-deposition machines (ALD).

So-called spatial-ALD involves sending the wafers forwards and backwardsdown a line of alternative precursor gas chambers to rapidly build upmonolayers of material.

FIG. 20 illustrates electric field distributions in the bi-directionaldevice. The two lower pictures are the electric fields modelled withSilvaco Atlas tool of +1400V and −1400V applied between CE1 and CE2terminals with BASE at 0 v. The simulation has cylindrical symmetryaround the X=0 line. When the simulation is run without thedouble-positive bevel, breakdown voltage occurs below 900V due to theelectric field exceeding 2e5 V/cm at the surface of the silicon.

FIG. 21 illustrates the doping concentrations in the bi-directional BJTdevice. The doping concentrations are taken in the Y direction in a linestraight-through the wafer, missing the BASE region. The right-handtrace is the doping profile at the BASE region.

Base Resistors On-Die.

FIG. 22 illustrates an array 2200 of CE1 stripes. On-die variations ofdoping levels, junction thicknesses and recombination lifetimes canprevent even a monolithic device acting as a single ideal device evenwhen the arrays of CE1 stripes are hardwired together. Instead it willact as a parallel collection of slightly mismatched transistors. Thismismatching can result in current crowding in certain regions of thedevice, hot-spots and generally lower than calculated performance. Forthe BJT type devices operating with a voltage mode base-drive, a 2:1mismatch of base current between any two stripes can arise from about a25 mV (kT/Q) offset in locally-process-dependent BASE voltage of the twostripes.

To reduce this effect, series resistance can be added to the BASE ofeach sub-device in the parallel array. This has the effect of moreequally sharing the common BASE input drive current amongst thesub-devices despite VBASE differences.

The required series resistance can be obtained by use of the P+ diffusedregions which themselves are shallow and quite resistive. FIG. 22illustrates how an air-bridge can be created by defining a narrow sliverof SiO₂ which upon etching is fully undercut and left suspended inmid-air.

SiO₂ Air-bridge concept can also be used on the JFET-base device inorder to link up the multiple bases electrically to one another.

Slab Inductor for Bootstrap System.

It is known that adding bootstrap circuitry gets over one of the olderdrawbacks of BJT technology, namely that a not-insignificant currentdrive has to be found to drive the BASE of the device. Thyristors andTriacs although bipolar technology, actually self-power the BASEcurrents from the through-currents (inherent 0.7 to 1V drop however),but the end user does not have to create a continuous current drive.Therefore Thyristors and Triacs are still very popular for AC powerswitching applications.

FIG. 23 illustrates a solid state relay module 2300 including a ‘slab’type inductor for bootstrap DC-DC. It is effectively a 1-turn ‘E’ core.The centre plate 2310 is twice the thickness of the top and bottomplates 2315, 2320. Flux is split 2-ways exactly like a standard E core.An air-gap default of the copper-foil thickness is apparent to allowhigh current operation without saturation.

The complete module 2300 is an intelligent power switch/relay/fuse ableto power itself from the through-current of the switching element whenon, and from leakage current when off.

Bi-Directional BJT Device Theory of Operation and Drive Systems.

Operation of the different regions of the device follows the principlesof bipolar junction transistors and/or junction field effecttransistors. Primary conduction path is formed when minority-carrierinjection causes conductivity modulation of an otherwise lowly-doped,voltage-sustaining, “bulk” region. This path is similar to that in astandard Thyristor and is therefore well proven. In terms ofconductivity it is preferable to use a P-type semiconductor for the bulkwhere the minority carriers injected are electrons which have at least2× higher mobility and diffusivity versus holes. Another reason forusing P-type semiconductors is that solar-cell P-type wafers, whichfeature an optimised high minority-carrier lifetime, are available atextremely low cost.

For the highest voltage operation (typically >2 kV) an N-typeconductivity-modulated region is commonly used due the availability ofnuclear-radiation doped silicon (radiation mutates silicon intophosphorous at a very well controlled rate).

A driver circuit is described later which supports both P-bulk andN-bulk devices and has facilities for bootstrap.

Description of the Operation:

FIG. 24 illustrates the off-state and on-state operations of thebi-directional BJT device in accordance with the present invention. Thediagrams attempt to show a coarse-grained doping level using circles torepresent electrons (filled) and holes (unfilled). The same symbols areused to show moving carriers during conduction. The diagrams show a halfof a device symmetrical about the X=0 line. In practice many ‘stripes’are arranged in parallel to form a large device. All diagrams are forP-bulk devices. N-bulk devices would have all voltage polaritiesreversed and N/P doped regions reversed.

Off State Operation of a BJT PNP-Type Base Device

FIG. 24( a) is an illustration of a bi-directional BJT device (BJT PNPbase) with zero bias on all the terminals (CE1, CE2 and BASE). A smalldepletion region 2405 exists around each junction of the device.

FIG. 24( b) is an illustration of a bi-directional BJT device (BJT PNPbase) with a large positive voltage on CE2 with the other two terminalsstill at zero volts. A large depletion region exists between the CE2 andthe bulk, drift region.

FIG. 24( c) is an illustration of a bi-directional BJT device (BJT PNPbase) with a large negative voltage on CE2 with the other two terminalsstill at zero volts. The area of the sunken base region 2415 is muchsmaller than the CE1 region 2420 with the N+ doping and this dopingcombined with the N− doping under the base helps to ensure that the P−bulk region 2425 is fully depleted giving much the same voltagebreakdown characteristic as a uniformly N++ top region would give.

On-State Operation of a BJT PNP-Type Base Device:

FIG. 24( d) is an illustration of a bi-directional BJT device (BJT PNPbase) with CE2 at +0.1V, CE1 at 0V and BASE at +0.6V. The main currentflows when the device turns on under a condition where CE2 terminalwhich is slightly +Ve (e.g. +0.1V). This represents the condition afterfully switching of a load has been achieved (prior to this there couldhave been over 1000 volts over the switch) and now just an ‘ohmic’characteristic is seen across the switching terminals of the device. Inthe PNP base region 2415, where the N part is part of the CE1 terminal,holes are injected and diffuse through into the P-bulk region 2425 ofthe main conduction NPN region making a low-voltage-drop connectionbetween BASE 2415 and P-Bulk 2425. Although both CE1/P-bulk andCE2/P-bulk junctions become forward biased, the larger forward biasexists to CE1/P-bulk (0.6V applied verses 0.5V applied over theCE2|P-bulk) and so more electrons diffuse into the P-bulk from the CE1end than the CE2 end. Also, more holes drift to that end of the deviceto provide the ‘base’ (recombination) current of the main NPNtransistor.

FIG. 24( e) is an illustration of a bi-directional BJT device (BJT PNPbase) with CE2 at −0.1V. This is generally the case after fullyswitching a negative voltage to a load. In this case a nominal 0.6V Vbeis achieved with only 0.5V of base voltage. The extra 0.1V is providedby the −0.1V CE2 voltage. For the reasons given previously and as seenin FIG. 24( e), hole current flows more down the CE2 where electrons arebeing injected more rapidly.

Note on Doping:

For efficient operation of the integrated two-transistor system it isuseful that the built-in voltage of the PNP base/emitter junction ishigher than the NPN base/emitter junction of the main input transistor.When this is done the PNP acts as a switch whose emitter hole currentmostly runs usefully to the bulk region 2425 instead of throughdiode-action to CE1. The built-in voltage depends on the dopingaccording to the following equation.

$V_{builtin}:={V_{t}{\ln \left( \frac{N_{a} \cdot N_{d}}{N_{i}^{2}} \right)}}$

Here Ni is intrinsic carrier concentration at temperature and Vt=k*T/q.This is usually the case when the doping of the bulk region is very lowto support a high voltage but it sets a minimum doping requirement forthe base 2415 of the PNP (and hence a maximum etch depth of thisfeature).

Operation of a JFET-Type Base Device

It should be stated first of all that the JFET input circuit does notgive the device a high-input impedance. This is because the JFEToperates in a ‘common-gate’ mode where the input terminal of thebi-directional (T2) device is effectively the source of the JFET.

Off-State Operation of a JFET-Type Base Device:

FIG. 25 (a) is an illustration of a bi-directional BJT device (JFET typebase) with a zero voltage condition. The N+ and N regions abutting theBASE will fully deplete the short vertical channel region 2510 of holes.Construction of this normally-off JFET type base normally requires verytight control of doping and geometry. Even the maximally doped N+ regionis only able to deplete around 3 microns of a typically doped P− bulkmaterial. This means that the vertical JFET channel 2510 will tend to beless than 6 microns in the X direction (assuming N+ is on both sides ofthe channel).

In FIG. 25 (b) the channel is still depleted when BASE=0V but in thiscase this may not have an effect since the high voltage drop issupported at the CE2 end 2520 of the device and there is no tendency forcurrent flow in the channel 2510 of the JFET type base.

FIG. 25 (c) shows that the CE1 and JFET depletion regions are joinedtogether. As the JFET region is only a very small area relative to CE1,there is negligible loss of ability to form a high voltage withstandingdepletion region in the usual way.

On-State Operation of a JFET-Type Base Device:

FIG. 25( d) and FIG. 25( e) show the on-state operation which is verysimilar to the PNP base type device as illustrated in FIGS. 24 (d) and(e). The only difference here is that holes flow directly from the BASEto the bulk whenever BASE is higher than the pinch-off voltage of theJFET that was formed. Holes do not need to diffuse through a region ofopposite doping as was the case for the PNP-type base in FIG. 24. Anadvantage of the JFET type base over the PNP type base input is thatnone of the BASE current of the T2 device is lost to base current of thePNP input stage which saves about 5% of the input drive current. Afurther advantage comes from the fact that the pinch-off voltage atabout 0.3V is generally less than 1×Vbe and could allow a driver circuitto remove charge from a saturated device without a reverse-biasedjunction preventing it.

Improved Bootstrap Circuitry

FIG. 26 illustrates improved bootstrap circuitry in accordance with thepresent invention. Unlike a standard Thyristor or Triac, the T2 devicesdo not have a mechanism for self-sustaining turn on. For Thyristor orTriac this mechanism comes at the price of approximately 1 W of loss peramp of switched current. The T2 device can reduce this by a factor ofaround 10 by exploiting the inherent current-gain of the BJT structure.However this may use additional circuitry.

FIG. 26 A, B illustrate an arrangement of switches which operate in twophases to multiple a low voltage at CE1 of typically <+0.1V to VDDvoltage of typically +1V. The MOSFET switches (Q1, Q2) are controlled bya PWM controller within a microcontroller. Q1 switches to charge theinductor 2610 with flux with C1 acting as input filter capacitor. Q2switches the positive boost current to VDD and onto C2. Voltage boost isset by the duty cycle.

FIG. 26 C, D illustrate arrangements in which CE1 is negative (as itwould be when an AC load current reverses). The current build-up in theinductor is of the opposite polarity. When Q1 disconnects in this case,Q3 is switched to VEE to generate a negative voltage of typically −1V. Adifferent switch Q3 steers the negative boost to C3. There is a standbymode where this DC-DC converter can be shutdown to save power.

FIG. 26 F) is a self-explanatory charge pump circuit. This canbi-directionally couple +VDD to −VEE transferring energy from one railto the other under the microcontroller command. This means that +1V and−1V (2V total) is available to power the microcontroller whether or notthe T2 device is switching is positive, negative or AC.

Drive Circuit Improvement

FIG. 26 G) is a base drive circuit which works with VDD, VEE and GND andis again driven by a PWM within the micro-controlled chip. This can becontrolled to produce a positive base current for a NPN type T2 device(P-bulk region) or a negative base current which would drive a PNP typeT2 device (N-bulk region). In discontinuous mode, the PWM ratio sets theBASE current.

At 100% PWM the maximum voltage available to the base is either +VDD or−VEE.

Initial Bootstrap Circuit

Prior to the Inductive DC-DC switch on, and prior to the switch on ofthe charge pump the circuit must accumulate enough energy to be able tosuccessfully start-up. FIG. 26 E) illustrates a circuit which can takeeither positive or negative leakage current and store it on VDD (C2) orVEE(C3) via the diodes D1 or D2. Circuitry inside the control chip e.g.low frequency microcontroller mode, can start up on just 1.2V fromeither VDD or VEE.

Providing that C2 and/or C3 are large enough there would be sufficientenergy for a self-sustaining boot up of DC-DC and charge pumps when theT2 device is commanded to turn on. Until a command to turn on thedevice, an ultra-low power standby mode can be maintained which suppliedonly from leakage. If the inherent T2 leakage current (thermallygenerated semiconductor leakage) is present, then a bleed resistor inparallel (shown) can ensure sufficient current. Q9 is an NMOS switchwhich is able to isolate capacitor C1 during the start-up, since C1might be of such a high value that during 60 Hz AC operation its voltagedoes not swing sufficiently to allow the diodes D1, D2 to operate.

Controller Chip Construction

Many of the MOS switches used in bootstrapping operate at high current(same current as going through the T2 device) but because they are lowvoltage switches they are fabricated on the same die using the samedeep-sub-micron transistors as the microcontroller. “On” resistance forNMOS FETs built on a 0.15 u process are approximately 0.0005 Ohm*mm². Ittherefore only needs several mm² of die area to implement those MOSswitches.

T2 Devices as being a Replacement of IGBTs

Background

This document expands on the driver circuits for the T2 device(previously called IBT in the text—particularly but not exclusively tothe JFET input transistor device). It notes that a re-think is neededfor the role of current-mode transistors in modern electronics. It alsolooks at how to make a functional equivalent to the ubiquitous IGBTtransistor at less cost and with better performance.

Comparing Current-Mode to Voltage-Mode Drive of Power Semiconductors inGeneral: Voltage Mode Gate Drive:

MOSFETs and IGBTs have a voltage-mode on/off control where a voltagesufficiently high (10V typ.) on an insulated-gate causes an inversionchannel in the underlying lowly-doped semiconductor switching it fromoff to on. Once this voltage has been established on the gate then nomore current is needed to maintain the switch in the on-state. Intheory, the driver circuit need only be a low current circuit and issimple to make. In practice, when switching quickly, many amps ofcurrent are needed to allow fast charging of the gate for efficientturn-on and off of the device. The average gate power consumption isstill very low but even so, MOS devices require a high voltage (typ. 15Vrated) control driver process able to supply several amps of pulsedcurrent. Historically, when the power MOSFET and IGBTs appeared in the1970s, 1980s most control systems used +/−15V supplies so the highvoltage gate was not seen as a problem. Today it is rare to see analoguecontrol circuits operating above 5 volts and logic ICs operate at lessthan 3.3V.

Current-Mode Base Drive:

The bipolar junction transistor (BJT) is the archetypal current-modedevice which requires a current drive into its base to turn it on. Thevoltage level is low, generally less than 0.9V for a silicon device overtemperature. Initially the current goes to charge the junctioncapacitance but continuous current flow is required thereafter tomaintain the switch in the on condition. The continuous base currentresupplies the injected carriers which are lost due to recombination inthe device. The higher the ‘Beta’ current-gain of the transistor, thelower the continuous base current needs to be.

Base current power loss is often seen as non-negligible, yet it isgenerally overlooked that this reaps disproportionate gains inefficiency elsewhere in the device. For example a 25 A collector currentBJT with a Beta of 10, needs 2.5 A of base current to stay fully on.Assuming a base voltage of 0.75V this equates to 1.9 W of power. To makea fair comparison between various MOS and BJT type devices, (and in thebase of bootstrap base-current generation where this base power is notseen externally), it is useful to consider this loss as an effectiveaddition to the ‘On’ resistance of the devices. The example loss justmentioned equates to 3 mOhms of on resistance which for a high-voltageMOSFET is an impossibly low figure to match.

For efficient operation, a BJT device should be driven with‘just-enough’ base current to maintain the collector current at low Vce(sat). With the technology available in the 1970s, base drive circuitsto achieve dynamic control of saturation were bulky, expensive andnon-optimal. Therefore BJTs were quickly displaced by power MOSFETs atleast in the low voltage application area.

Today however where a single chip microcontroller is able to fullycontrol a BJT in real-time (32-bit microprocessor including FLASH, RAM,12-bit ADC, Quad PWM, Serial ports can be bought for ˜$0.50) thesituation is completely different.

Furthermore, low voltage, high current systems such are now common—atypical server microprocessor is powered over PCB traces at 150 amps at1 volt, and inductor based DCDC modules to supply these needs are small(recently this entire system including inductors are integrated on-chipon the Intel Haswell microprocessor).

Advantages of Current-Mode Low Voltage Drive

-   -   can be implemented in standard logic process with PWM, adc and        drivers (low voltage ˜0.9V maximum needed for a P/N junction in        silicon)    -   possible to integrate the with microprocessor with driver for        system-on-chip    -   reverse base current can turn off devices quicker than without        (compare to IGBT)    -   Adiabatic recovery of a much of base charge (inductor versions)

Inductive Base Drive Circuits for Current-Mode Power Transistors

Previously the benefits of having multiple base current balancingresistors have been discussed to ensure matching of the BJT fingers (seethe description with reference to FIG. 22). Should there be a problemwith uniformity of silicon characteristics within the die of a singledevice. The same balancing effect can be achieved by modification of thepreviously described inductive buck-mode PWM driver to split the outputinto multiple inductance branches of the base current driver. This isshown in FIG. 27 a for matching on-die but could also be used to matchmultiple die which are operated in parallel. Again this feature isoptional. An advantage of using inductors to match currents is thatthere are lower losses than using resistors. The variation of currentfrom one device to the next is proportional to the difference betweenVBE and the peak drive voltage to the inductor.

During the ‘On’ period, the average base current can be controlled usinga discontinuous-current inductor drive. Base current is proportional tothe PWM_ratiô2. Discontinuous operation occurs when the off-time is longenough for the inductor current to drop to zero before the next positivepulse. The discontinuous nature will not cause switch-off of the T2device if the PWM frequency is sufficiently high. When the PWM period isshort the amount of charge given in the PWM period is much less than thesaturation charge accumulated in the BJT junctions of the T2 devicewhich act as a capacitive filter to the base current.

FIG. 28 is a transfer curve of current vs. PWM value (0-255 range) forone path showing that discontinuous current drive is highly non-linear.

Advantageously, if the PWM inductive paths are split and independent, itgives an opportunity to add multiphase operation.

FIG. 27 b suggests that the same PWM duty cycle is given to each of thepaths or advantageously the waveform of each path may also bephase-offset between each other to help to reduce switching noise andreduce ripple current.

Using ADCs, the microcontroller can be aware of and command the positiveor negative current build-up (turn-on, turn-off respectively) by holdingthe output positive or negative until the desired current is calculatedor measured to exist. Or it can set a level which is good for continuousconduction—just sufficient base current to keep the voltage drop overthe device at the ideal point based on saturation detection ADC resultfrom the CE2.

Fast Turn on Scheme with Inductors:

One downside of series-inductive base drive is that it limits therate-of-rise of base current (by definition of inductance) and thismight increase the turn-on time of the power transistor and thereforeits switching losses. For most applications, and with say a 1 MHz+PWMfrequency and low value base inductors (<1 uH), the fact that the mainT2 junctions will not turn on strongly and immediately (until a P/Njunction charge has been established) lets the base inductive currentbuild up strongly so that it ultimately pushes through the loss-dominantturn-on stage very quickly (less than 2 uS). If higher turn-on speedsare desired, then the IC transistor driver topology shown in FIG. 27A or27F can sidestep the normal delay in base current build-up using apre-charge scheme. Since the driver IC is intelligent and digital, andthe time of the positive drive current is known in advance, a pre-chargeperiod can be deployed to get the inductor current up to the requiredlevel before applying the current to the base.

FIG. 27 C has the turn-on scheme waveforms. The current is applied tothe base only after it has fully ramped to the desired level.

Turn Off:

Turn-off can be done using opposite-voltage opposite-current drive.Storage time (time before all the charge is removed with the transistorstill conducting well) allows the current build-up without the need forspecial measures as used to turn it on. But, a more complex drive schemecould be envisioned where the inductor is electrically ‘swapped’ [DPDTMOS switches] to instantly reverse the base current keeping itsmagnitude and this would afford a very quick turn off.

Alternatively, an intelligent driver is able to reduce base currentahead of the known-time that the transistor is to be switched off,letting it come out of saturation, and this will speed the transistorspassage through the turn off region reducing toff losses.

Adiabaticity:

During turn-on and turn off much of the charge transferred to the baseis adiabatically stored/recovered by the inductor/supply capacitorsystems. This allows use of very large +ve, −ve currents to be deployedduring turn-on, turn-off of the device giving high switching speed andlowest switching losses.

Capacitive Speed Up of On/Off:

Of course capacitor-base conventional BJT drive circuits could be used.

Fastest Digital on/Off Current-Mode Driver for High Speed IGBT-TypeApplications

FIG. 29 is a schematic diagram of a digital current mode driver 2900.Multiples of these can drive base fingers or die. The example is for anN-bulk T2 device such as the NIGBT structure (described later) where themain transistor is PNP and the base is either an NPN or N-JFET needing a−Ve polarity current to turn on.

The power source VEE_(adj) typically −0.9V comes from a DC-DC convertermade using a single inductor in the normal way (could be the bootstrapDC-DC) and is programmable by the VDAC set sufficient to drive the Vbeof the highest-voltage finger plus some margin to account for FETvoltage drop. Feedback from the ADC information can be used to fine tunethis voltage which could be set higher just prior to turn-on of the T2device for speedup.

This scheme is non-adiabatic but the current build-up does not depend oninductor ramp-times.

Fully digital ‘DAC’ control of individual BASE current outputs.Resistance between the BASE (VBE) of the device and the slightly greaterVEE_(adj) (typically a 0.1 to 0.2V difference) sets the BASE current.

Digital size-weighted FETs form an array (3 bits shown, could be 8 ormore) making effective binary setting of this resistance (formed of theFET ‘on’ resistance) connecting VEE_(adj) to the BASE of the T2 devicefinger(s) and gives an overall DAC function for current.

Multiple units of the above can independently drive multiple basefingers and/or multiple die.

Turn-off of the T2 device is by the JFETs in this example. Unlike astandard IGBT it is possible with the T2 device (especially the JFETversion which has ˜0.3V barrier potential) to actively remove chargefrom within the junctions and the bulk of the device during turn off.Advantageously this shortens the turn-off time and reduces switch-offenergy loss.

Because a low-voltage, current-mode base drive is used, the system canbe implemented fully within in any standard deep submicron CMOS ASIC. Toget an idea of the area needed, a 0.18 u CMOS process (1.8 volt) hasNFET on resistance of less than 0.001 ohm*mm2. For example, a 100 A basecurrent driver requires only 1 mm² of silicon NFET area in the ASIC toswitch this current (in practice getting the current in and out of thedie takes up more space as does the buffers but the cost is still low).

Digital control permits ‘on-the-fly’ adjustment of drive at multiplepoints in the output power cycle. For example, at turn-on a very highcurrent could be given and during the cycle the base current can beregularly trimmed to keep the T2 transistor at the optimum point ofsaturation. Close to the end of the on-period, the current could bereduced to minimise the stored charge prior to finally an oppositecurrent used to turn the device off.

A more advanced version could programmatically trim out mismatches inthe various characteristics of large transistors through pre-programmedoffset and gain currents between finger drivers determined bycalibration. This would give more uniform current density in the T2devices.

It may be more desirable to mount the ASIC on top of the T2 die and wirebond between the die.

Combination Drivers

One or more of the approaches mentioned can be combined for optimumeffect.

It might be possible to integrate some or all of the control electronicsonto the T2 transistor die.

I2 Device: A Combination of T2 Device and Driver Replaces IGBT with aNIGBT “Non-Insulated-Gate-Bipolar-Transistor

Review of Structure and Operation of the Standard Planar IGBT

FIG. 30 (a) illustrates a cross-section of a standard planar IGBT. Whenthe gate terminal is taken to typical +10V, an inversion sheet forms onthe P region under the gate giving an N-type connection from the N+emitter to the N− base region. This feature does not in itself injectminority-carriers which instead are injected from the collector P+region when the collector is positive. This has three sub-optimaleffects. 1) The On-state voltage can never be less than 1 diode drop. 2)Although the device is otherwise capable of AC voltage blocking, itcannot be switched on for the reverse polarity direction. 3) There is noexternally available contact to the N− region which might be used toremove charge to effect a rapid switch-off.

The equivalent circuit for an IGBT is often represented as a PNPtransistor with its base switched by a NMOS transistor. This is a goodmodel for those devices where the PNP beta is designed to be relativelyhigh (>4). In those devices the ‘base’ current flows as electronsthrough the NMOS while a much larger current of holes diffuses from C toE as minority-carriers in the base.

Another model that of a PIN diode in series with an NMOS devices isappropriate where the beta of the PNP<˜2. In these cases, recombinationis so high that most injected holes do not make it from C to E but arepresent in enough numbers to reduce the resistance of the N− layer likea PIN diode. Current in the NMOS is a higher proportion of the totalswitched current in this case with less aid from the transistor Betaaction.

Structure of the I2 Devices Used to Replace IGBT

A N− bulk version of a I2 device see FIG. 30B made with either a BJTbase (in this case an NPN base) or a JFET base I2 device each operate ina similar way by injecting electrons into the N− base and are justreverse-doped versions of I2 structures already described. Polaritiesand current flow directions are reversed relative to a P− bulk T2device.

Whereas in an IGBT, J-FET regions are considered ‘parasitic’ and lead tothe development on the trench IGBT to avoid them; in the I2 JFET-basedevice the JFET region is encouraged and essential for the off-state. Inthe on-state the depleted channel disappears to be replaced with aminority-carrier-injected channel orders of magnitude higherconductivity than a MOS inversion channel.

Operationally, the main current-carrying PNP transistor is virtuallyidentical to the IGBT PNP mode (and for that matter to that of aThyristor), but in this case it can operate without a built-in diodedrop in the C to E (CE1 to CE2) main conduction path.

The gate/base arrangement is very different. All of the insulating gatematerial and contact of an IGBT are no longer present and there is no N+region connecting to the E any longer. This eliminates the possibilityof NPNP latch-up because the N+ contact is not to a potential orpermanent low impedance capable of sustaining latch-up.

For these N− bulk I2 devices, a negative base voltage of −VBE at currentof Ibase gives BETA×Ibase of current through C to E terminals. In casesof low BETA of the PNP, the PIN action will also be present (like aPIN-mode IGBT) and the base will have to take most of the switchingcurrent of the device. In the limit, with if there is no PNP action, theefficiency of the I2 device will not much exceed the IGBT except for theelimination of the MOS resistive channel voltage drop of an IGBT.

Punch-Through/Non-Punch-Through/Electron Irradiation Options:

The same set of design optimisations used for IGBTs such aspunch-through/N+buffer layer can be applied to the I2 devices making itasymmetric and not able to support (much) reverse voltage.

The region label BL can be N−, N+ or something in between, exactly likestandard IGBT processing for Field-Stop, Soft-Punch-Through,Controlled-Punch-Through, Light-Punch through etc. Electron irradiationcan be applied to the T2 type devices as it can to traditional IGBTs forlifetime control.

Transparent Emitters/Collectors:

There also seems to be no reason that a transparent emitter or collectorcould not be achieved also if desired. This occurs where the doping isso shallow that the carriers can pass right through and recombine on thehigh-surface recombination velocity metal contacts.

Inherent Inverse Diode Possibility

For any version of the I2 device the AC conduction ability of thestructure can give ‘for free’ a programmable inverse-parallel diodefeature (whereas the IGBT has no way of turning on the thick-base BJTwhen the collector goes negative). When the I2 driver IC detects anegative collector voltage it can optionally drive the base current inwhich case holes will be pulled out of the top P+ junction turning onthe transistor and clamping the negative excursion. This clamping actionis ohmic (saturating) so can be as low as 0.1V at moderate currentdensities.

I2 “NIGBT” Advantages Over IGBT

-   -   much simpler fabrication process (does not need oxide, fewer        masks, less diffusions).    -   much more efficient    -   free of the P/N diode drop    -   ‘ohmic’ like saturation ‘on’ resistance, or non-saturated under        program control.    -   Direct control of the internal charge of the main base junctions    -   Possibility to do fast switch off by actively removing charge        from the base    -   AC or DC switching    -   inherent anti-parallel diode action (ohmic) with special driver        (whereas IGBT needs an external diode or extra processing steps        to fabricate one)

Example Conversion of IGBT to NIGBT I2 Device

To illustrate the process of converting an IGBT to a I2 device, we tookthe planar IGBT example distributed with Silvaco's Atlas TCADdrift-diffusion simulator.

See http://www.silvaco.com/examples/tcad/section40/example4/ for theexample.

With the same doping and geometry, stripping out the MOS part andrearranging the terminals, and contacts, an I2 version reduced thesimulated on-voltage from 1.75V to 0.15V at the cost of (equivalent VCEloss) 0.5V due to base current. This cost dropped to 0.2V equivalentwhen the lifetime was increased from 1 uS to 5 uS (still well withinstandard CZ wafer specifications).

The overall benefit is a 2.5-5× reduction of conduction losses,improvement in switching performance, a driver that can integrated on aCMOS chip, and 2 fewer manufacturing masking steps.

AC Current Conduction Paths

When considering the AC operation of an I2 device, FIG. 31 helps tovisualise the main current paths in each direction of operation. In thiscircuit, the Beta of the device overall device is 10.

(Note: Beta=Alpha/(1−Alpha); Alpha=Beta/(1+Beta))

The I2 device can be further simplified for a DC application in whichcase it does not require a second transistor element such as a JFET. SeeFIG. 30 C in which a lateral version of the I2 device is illustrated).The construction techniques mentioned before are usable but the etchedbase feature could be deep enough to cut right through the CE1phosphorous diffusion.

Matrix Converter Application

Possibly the single largest application for medium voltage, low/mediumfrequency power switching is in variable-frequency motor driveapplications—currently $18 Bn per year for these inverter drives.

Roughly 50% of all electrical power used in the world goes into electricmotor applications where a variable frequency drive can reduce this byaround 18% typically (sourcehttp://en.wikipedia.org/wiki/Variable-frequency_drive).

Currently, motor applications which have been converted tovariable-frequency operation are claimed to save over 1 billion tonnesof carbon per year in emissions.

Variable-frequency drives represent 3% of the installed base and areonly fitted to 40% of new motors entering the market (the rest aredirectly connected to 3 phase line power) leaving a huge untappedpotential carbon saving.

The biggest reason to the low adoption rate is cost of the inverterelectronics and secondary problems caused by increased input harmonicpower from standard switching electronics such as IGBT AC-DC-AC baseddrives and reduced reliability due to DC bus capacitor failure.

The well-known and extensively researched Matrix Converter AC-ACconverter can circumvent all the known issues and has the followingadvantages for motor drive applications.

-   -   Very low input harmonics    -   Good power factor    -   High reliability by exclusion of DC capacitors    -   Bi-directional energy flow—breaking, regeneration possible        (Matrix converter topology is actually a general AC-AC power        conversion topology not restricted to motor driving).

FIG. 32 shows a Matrix converter system topology for very low cost andhigh reliability using I2 devices and driver techniques already outlinedin this document. An array of 3×3 AC switches is fundamental to theoperation of direct Matrix converters and the I2 devices are suitablefor this application.

When grouped in arrays of 3×I2-devices they can be driven locally by3-output driver IC containing 3 copies of any of the various drivecircuits discussed. A central control circuit can coordinate theswitching of the 3 banks of triple-devices.

Device Naming Conventions:

T2 device—non-punch-through design with almost symmetrical forward andreverse conducting and voltage withstand ability. It has a 2 transistorstructure and also has high voltage, high current main BJT powerstructure—NPN or PNP. In addition, T2 devices have an input basetransistor of either JFET or BJT construction, for example, T2 NJ—an NPNwith a JFET input transistor, and T2 PB—an PNP with a bipolar inputtransistor. It will be appreciated that the T2 devices are previouslyreferred to as an IBT device, particularly but not exclusively, designedfor an IBT JFET transistor version.

I2 device—punch-through, or field-stop device to compete with IGBT andalso known as “NIGBT” (non-insulated gate bipolar transistor). Thisdevice is the same as the T2 but the addition of a field-stop or otherburied layer reduces the voltage withstand ability in one direction toaround −20V while the current capability of both directions ispreserved. The I2 device have a two transistor structure—same options asT2 (see above). I2 devices operate in “Emitter follower” configurationsand are useful even when the main transistor Beta drops below unity.

B2 device—usually but not necessarily punch-through, or field-stopdevice to compete with IGBT so another kind of “NIGBT”. Like I2 devicethe voltage withstand ability is low but now—0.6V max. 1 transistorstructure—a vertical BJT. When operated with correct driver it has aninherent reverse diode which comes ‘for free’ and can be conductivitymodulated by reverse saturation by the base for very-low forwardvoltage. B2 devices operate in common-emitter mode.

Field-Stop (FS) Devices.

A Field-stop is no different in principle to a punch-through designdiscussed before where there is a buried layer of extra-high dopingwithin the voltage sustaining base (drift) region of the device.

(See FIG. 30, where BL is the buried layer). The field-stop is oftenjust thicker and less highly doped than a buried layer but to the samepurpose. Referring to an NPN (P-base) device, when the reverse bias ofthe high voltage junction exceeds a certain limit, the depletion regionfully extends through the P-base layer 3015 and only the higher P-dopedField-stop layer 3010 prevents the depletion region from reaching theEmitter N+ diffusion 3025 and causing breakdown. The device has a muchhigher breakdown voltage for a given die thickness in its preferreddirection but is no longer a symmetrical AC switch, since in the reversedirection the field stop layer's doping has the effect of dropping thebreakdown voltage considerably. Nevertheless DC switches are preferredin many applications.

The effect of the field-stop on the T2, I2 and B2 devices is similar tothat in an IGBT or PIN diode where punch-through operation also helps tospeed up the device turn-off without needing additional lifetime controlmeasures.

FIG. 30D illustrates a schematic diagram of a B2 device including avertical thick-base BJT with a Field-Stop layer 3010. This design is afully planar version but anisotropic-etch versions are also possible. Ahalf-section of 20 u width is shown.

The field-stop layer 3010 for an NPN B2 device will be a P diffusion oftypically 9 u thick and doping levels of around 1e15 to 1e16/cm³ can beformed with either Boron, Aluminium or Gallium impurities, the lattertwo having fast diffusion speed and can possibly be co-doped (co-fired)with the phosphorous impurities for a single furnace diffusion processyielding the desired junctions. A suitable reference for this processcan be U.S. Pat. No. 3,681,155A.

SiO₂ can be used as the mask material against phosphorous to form thejunction patterns.

FIG. 30 G illustrates a top view and bottom view of a full die madeaccording to the scheme (before metallisation). Field rings on thebottom side are for edge-termination and reduce the electric fields atthe border of the die (unlike an IGBT, in the B2 device, the depletionregion starts at the collector (bottom) 3020 side).

Because the device operates in common-emitter mode, the Beta graph ofFIG. 30F is particularly important because at Beta less than ˜0.75, thedevice becomes less efficient than an IGBT. At the intended operatingpoint beta is around 2.5 with an effective Vce (sat) effective of ˜0.5volts and even lower at lower current densities.

Driver IC Improvements

It will be noted that the description here refers to multiple basefingers of a single power transistor die but applies just as well tofingers spread over multiple power transistor die. It assumes alwaysthat there is a microcontroller or other sophisticated digital machineacting through a control algorithm to make PWM and other outputs basedon information received from ADC channels (See for example FIG. 2B).Also PMOS transistors are used for positive switching which could besubstituted by NMOS with the appropriate bootstrap or auxiliary powercircuits. The drivers here are useful for all kind of current-inputdevices such as Silicon-Carbide BJT, GTO Thyrisors, Standard BJT devicesand BMJFETs (bipolar-mode JFETs)

Inductive Base Drivers

FIG. 27 A, B, C illustrate drivers which are able to drive NPN or PNPdevices because of a +/− output stage;

FIG. 27 D, E are base finger drivers which can also be built withstandard CMOS low-cost process but simplified for one polarity drive orthe other, not both at the same time;

FIG. 27 F graphs the Spice simulation of the drivers and indicates thedesign features needed of the PWM mechanism to implement a flexible,efficient and very fast base drive response.

Dual PWM Per Base-Finger Driver

The key idea for a fast on/off current at the base is the dual-PWMsystem where the PWM1 is implemented with a fast low voltage on-chiptransistor (since it only sees Vbe) and gates the inductive current intothe base. It can also rapidly discharge the stored charge of the base.

PWM1 is just like a standard type PWM control signal for the transistor(although the PWM positions can be controlled in a specific way andindependent from one base finger to another as will be outlined below).

PWM2 is a faster-rate PWM which controls a synchronous buck-typearrangement on the inductor L1 which converts from a higher DC powersupply voltage to the Vbe voltage.

As indicated in the waveform, PWM2 can be used to pre-charge theinductor current within PWM1's off-time before it is gated onto the Baseoutputs at high speed and full strength.

During the ‘On’ period, the PWM2 can by dynamically adjusted e.g. fordynamic saturation control based on the instrumented readings of thetransistors operating conditions from ADC channels like FIG. 2B. Keepingthe transistor just within saturation prior to complete switch-off givesthe lowest E_(off) (energy off) figure.

FIG. 27 D, E have ADC take offs at the base finger itself and from acurrent sense resistor ahead of the base. This permits a measurement ofactual base current and base voltage.

Independent/Semi-Independent Multi-Finger PWM Drivers

Each finger-driver has its own PWM1 and PWM2 generator driven undersoftware control to control the On-Time, Off-Time independently of theother finger drivers. This ability can be used in software correct fornon-uniformity over large power-transistor die.

FIG. 33(B) shows a logic circuit which is able to re-time an existingPWM signal from say an existing PWM control IC. By tuning the final onand off position independently any mismatch between transistor regionscan be tuned out (mentioned before) but also any transistor Storage-timedelay, which increases the off time, distorting the PWM signal, can behidden with a corresponding extra delay to the turn-on position—thefinal switching waveform will be a delayed but overall faithfulreproduction of the PWM input. Most analogue feedback control ICs willnot notice this slight delay.

Combining Multi-Phase Drivers for Ripple Reduction on a Single Output:

FIG. 27 (G) gives another other option for multi-phase multi-finger basecurrent generators feeding into one terminal—good for standardtransistors e.g. silicon-carbide BJT devices. Multi-phase operation ofeach driver can smooth out the base current ripple as can be seen fromFIG. 27 (H) while still maintains the very fast On/Off base currentcontrol. It has the net effect of a higher PWM2 frequency butmaintaining the low losses of the lower frequency.

3D Driver/Power-Transistor Stack:

The physical construction shown in FIG. 33A which has a driver chippermanently attached to a power transistor (T2, I2, B2 or other) usingan interposer flex-PCB. Inductors, capacitors and other surface mountcomponents can be added. An intelligent driver IC is solder mounted tothe PCB from above flip-chip style (WLCSP (Wafer Level Chip ScalePackaging)) using bumping technique. The power-transistor device issolder mounted from underneath again using wafer-scale bumpingtechnique.

Other options could see the interposer made from a silicon substrate toeliminate any CTE (coefficient of thermal expansion) problems betweenthe die. Or, the main transistor die can have formed upon it additionalpatterned dielectric (e.g. polyimide)/metallisation layers toeffectively build an equivalent of the interposer directly on the powertransistor where the inductors and driver die could then be soldered.

On-Die Temperature Sensing of the Power Transistor at Multiple Sites:

Each base finger represents a PN junction and as such has a well definedforward-voltage (Vf) vs. temperature. Rather than taking the absolutevoltage measurement of Vbe which is one option, it is more accurate totake a Vf measurement at one current level and then another measurementat a different current level. The difference in readings varies withtemperature and eliminates a large unknown initial Vf voltage.

The inductor-driven base waveform (FIG. 27F) has within it varyingcurrent levels and so by sampling at intervals where base current ishigh and again where base current is lower using a different ADC channelfor each finger driven then a reasonable map of die temperatures isalways available to the control software. This is without needing to addspecific temperature sensors to the system or driver die (anotherpossibility).

FIG. 35 (A) is a small modification to a standard CMOS process tooptimise it for the role of driver especially of NPN versions of thepower transistor where most of the PWM conduction current is via NFETdevices to/from 0V. The N+ source diffusions have extensions deeper thannormal to connect directly to an N+ substrate which in turn can besolder bonded to the metal lid of FIG. 33.

The process modification is small—similar to a deep N-well procedurecommon on CMOS processes.

The resulting NMOS devices have fully vertical current conductionpaths—between substrate and the bond pads. This means that no lateralcurrent is taken through the normally thin metallisation of CMOS chipsso avoiding the issue of electro-migration which tends to limit currentto a few mA/micron width. Having the bond pads over the active siliconarea means they do not add to the active area consumed.

FIG. 35 (B) illustrates a simplified synchronous rectifier system forisolated power and data to/from driver IC using the CMOS chip. Phasemodulation of the edges of the AC switching by first the master (Driver)then the Slave (CMOS driver chip) can pass digital information in eachdirection along with power transfer.

Software and Firmware:

When the features described above are combined with a microcontrollerwith on-chip non-volatile (NV) memory there exists an opportunity tofinally fix several problems which have plagued minority-carrier devicessuch as GTO Thyristors and BJT devices in the past.

These problems include:—

-   -   Current crowding on large die leading to premature failure of        those parts of the transistor    -   with higher gain, higher temperature or higher carrier lifetime        than the mean.    -   Part-to-part variation of current gain—where replacement parts        do not function identically to the original.    -   Turn-off, dynamic breakdown hot-spots—leading to failures in GTO        Thyristors.

After manufacture of the Driver/Transistor stack, a calibration routinecan be performed on the complete device.

One such test the Open-Circuit Diode-test which gives an approximationto the minority carrier lifetime and can be done even without aspecialised test fixture. The driver can turn on a Vbe junction andsimply measure the rate of droop of voltage with the Vbe junction afterletting it float.

Results of the calibration are stored in non-volatile memory on thedriver IC e.g. Flash/OTP or fuse memory.

Calibration routines can performed in a dedicated test fixturesequentially exercising each of the finger driver channel and determineperformance of only that area of the power transistor die. As well ascalibration data, the non-volatile memory can hold defaults such astemperature trip points.

The calibration values are later used during normal operation of thedevice to compensate for the inherent variations within the powertransistor and external components (e.g. inductance variations) so thatfrom the end-users view every device operates correctly and nearlyidentically to any other device of the same type.

For example, if the driver chip software accepts a digital input forcurrent limit in mA it can use its own lookups of Beta scale factors ofeach finger driver for the X and Y region of the device (with 2D lookuptable), and by controlling On time and Off time at independently of eachPWM finger driver channel to meet the commanded current.

Current crowding on a large die will be reduced and inter-changeabilityis realised with easy series/parallel connections of devices.

Parameters which vary with temperature can be tuned out given thetemperature measurement from the particular region of the die and knownparametric variation of transistor parameters with temperature.

Timing variations over large die are known to cause dynamic breakdownfailures of power devices during turning off especially those which relyon long minority-carrier lifetime to maintain low switching losses. Forexample GTO Thyristors (PNP wide-base transistor at the core) haverelatively good turn-on characteristics but during switch off,variations of carrier-lifetime, resistivity and doping levels over thedie area during dynamic breakdown makes the turn-off process uneven.This can mean that finally a small region which happens to be last toswitch off must momentarily conducts all of the switched current and canburn out. This problem can be solved with the smart driver usingindependent pre-calibration timings applied via PWM for eachfinger-driver to offset for the turn-off time variations.

Finally with a digital interface there is the opportunity to hide theunavoidable Storage-Time parameter of BJT type devices where it takes aset amount of time to remove the charge from the base before the devicecan turn off. If the smart-driver accepts PWM turn-on and turn-off timesas digital words, the already-known storage-time can be subtracted fromthe given turn-off time to position the turn-off edge at the correctpoint in time.

Matrix Converter Boost System

The AC version of the T2 device seems ideal for Matrix converterapplications compared to standard DC-only switches. Only one bigdrawback remains for the Matrix converter motor drive in that it canonly create an output of approximately 86% of the input voltage. Thisstops the matrix converter working as a drop-in replacement for standardDC-bus 3-phase inverters.

A simple solution for this would be to boost the AC input voltage by 14%using auto-transformers.

However with 50 or 60 Hz transformers these would be bulky and eliminatemost of the advantages of the matrix converter. To address thislimitation a high-frequency AC-AC power converter is proposed in FIG.32. Unlike an AC->DC-AC based high frequency transformer, it hasinherent bi-directionality so it does not lose the inherent advantage ofa matrix converter with regards to braking and regeneration.

The AC-AC transformers are not large being only 14% of the power of thematrix inverter.

Conventional 3-Phase Motor Inverter

T2, I2 and B2 type devices can still be applied to reducing power lossesand cost in standard AC-DC-AC topologies like FIG. 36. Here the devicescan perform mains-voltage rectification and output switching ateffective forward voltages of around 0.25V reducing the power loss by afactor of around 4. The I2 device is particularly suited as a high-sideswitch and a single driver can activate many switches with a commonsupply rail—unlike NFET or IGBT high-side switching systems whichrequire separate floating supply and driver chip for each driver.

Low Leakage Relay Contact

Compared to electromechanical relays, the on-resistance of the T2switches can be made equally low by increasing the device area. Leakagecurrent can increase in proportion to the area of the device and evenwhen switched off there could milliamps of leakage—enough to causeproblems for some loads.

FIG. 34 proposes a solution using a shunt element Q2 device to bypassthis leakage current around the load when Q1 is switched off and whereQ1 is much lower on-resistance than the load resistance this solutionwill work well.

To take the leakage current to Pico-amp levels even with very lowimpedance loads, Q3 can be added and it turns on and off with Q1. Whenon it has to take the full load current but when off only has to supportunder 1V typically and so could in fact be built onto the driver chipwhere ultra-low on resistance FETs occupy only a few mm² (1.5 mOhm*mm in0.35 u CMOS).

Active Self-Powered Switching Devices and Circuits

These embodiments relate primarily but not exclusively to the driving ofBipolar Junction Transistor devices coupled to control circuits whichmay use BJT, MOS or other FETs. Descriptions here are generally furtherembodiments of ideas described in the embodiments above and especiallyrelevant background materials are FIG. 2C, FIG. 6, FIG. 13A, B, C, FIG.26, FIG. 27, FIG. 30 B,D. FIG. 33A, and FIG. 36, which are describedabove and which are applied often in relation to Circuit Breakers.

BJT devices require a Base current which is a fixed proportion of theCollector current to maintain saturated (low loss, ohmic) conductionmode. In this saturated conduction mode minority-carriers conduct bydiffusing from the Emitter to the Collector terminal with the Basesupplying carriers lost to recombination and at potential sufficient toovercome the built-in PN junction potential—typically 0.7V in silicon,and up to 3V for high-band-gap semiconductors such as Silicon-Carbide.

For a BJT transistor whose typical characteristic “Beta” [ratio ofcollector current achieved vs base current given] is 20, a circuitdesign point of 10:1 Collector:Base operating current might be chosen.This gives sufficient extra base current to account for part-to-partvariations and ensures the transistor will be fully saturated to a lowVCE (sat) voltage when turned on. In the example just given it is deemedthat the transistor is operating with a “Forced Beta” of 10.

In the literature there are very many application circuits to give aforced-beta for a transistor but generally these are not dynamic. If aBJT base is driven at fixed 1 A but the collector load varies from 1 Ato 10 A, only at 10 A collector-current is the forced-beta equal to 10.At lower loads the forced-beta is lower and power is being wasted in thebase (=Ibase*˜0.7V). If the load happens to exceed 10 A the transistormight be liable to turn off (insufficient base current) and could easilyoverheat as it VCE rises dramatically.

Historically, BJT devices began to fall out of favour in the 1990s, notbecause of the switching efficiency—which is unmatched amongsthigh-voltage devices—but because of the cost involved in providing a(relatively) high base current which has to dynamically adjust inproportion to the instantaneous collector current. This provedproblematic, especially compared to power MOSFET and IGBT devices whichhave no such requirements.

It is one object of the present embodiment to provide a simple andautomatic means of dynamic base drive of BJT-style devices.

One device of the BJT era which is still prevalent in the market is theThyristor. This device ‘scavenges’ Base currents (recombination losses)directly from the current in the circuit in which it operates and doesnot need dynamic base control. In a Thyristor the full load currentpasses through a PNPN junction stack, or equivalently a PNP/NPNcross-coupled-and-merged transistor (the NP junctions of the PNP aremerged with the NP junctions of the NPN transistor). The Base currentsare provided naturally from the larger flow of full current carriersthrough the device. Unfortunately the silicon thyristor must have avoltage drop of at least 0.7V in the full load-current path to overcomethe remaining PN junction voltage and this voltage drop is at the fullload current. Also once the thyristor is turned on, it is difficult toturn off since the base currents come directly from the through-current.

It is another object of the present embodiment to furnish base drivecurrent from the circuit current but overcome the usual 0.7V voltagedrop and a further goal to make non-latching devices possible.

More recent prior art devices which seek to ‘scavenge’ power from theload to maintain their turn-on are based on MOS technology which haveessentially zero gate current making it easy to use a charge pump orsimilar to extract operational gate voltage from across the terminals ofthe ‘closed’ switch. This approach however has three drawbacks:

1) It requires a disconnect-circuit to decouple the driver from thehigh-voltage side of the switch at switch-off time.

The present embodiment aims to eliminate this by using only low-voltage(<5 volt) devices and capacitor operating on the low-voltage side of themain transistor with no need to touch the high voltage side of the maintransistor at all.

2) Extracting driver power from a boosted version of the voltage overthe ‘closed’ power switch means that the switch can never be fullyturned-on for if it were, there would be no operating voltage with whichthe driver circuit can operate.

The present embodiment theoretically allows the main switch to turn onto zero-volts and no control needs to be applied to maintain aparticular switch voltage drop.

3) MOSFET devices are very large, inefficient and/or expensive whendesigned for high voltage operation.

Bipolar high-voltage device technology is the only option to compete oncost in most applications.

Embodiments Diode D2-Device

The features of the method are described by way of an example, thesimplest embodiment being that of an active rectifier. Externally thisis a two terminal device and acts as a low-forward-voltage, high reverseblocking voltage diode. Four such devices can form a bridge-rectifier.

FIG. 37 shows the circuit and structures for producing a 2-terminal‘diode’ device with a forward voltage of the order of 0.1V and a reverseblocking voltage of >1000V at very low leakage current. This performancecontrasts with traditional high voltage silicon rectifier diodes whichexhibit a forward voltage between 0.75V to 1V and cause a significantpower loss especially on 120 Vac mains rectification applications (2diodes conducting) of around 2 W per amp which equates in monetary termsto $2 per amp, per annum wasted on electricity per bridge-rectifier.

Silicon Schottky-type rectifiers are not useful at high reverse voltagebecause a) the reverse leakage current is very high and b) a Schottkydevice designed for high reverse voltage unavoidably results in a highforward voltage.

FIG. 37B shows the operating principle given in context of a ‘diode’ butequally applicable when driving true transistors such as an I2 device(as described in relation to FIG. 30 above).

An inductor L1, a capacitor C1, main switching transistor Q1 arecontrolled by the operation of two switches SWA and SWB operating athigh speed of the order >10 KHz<100 MHz in two phases called Phase A andPhase B. There is also a period where neither switch is activated calledPhase C. The numerical example here has F=2 MHz, a diode through currentof 10 A t(Phase A)=450 nS:t(Phase B)=50 nS ratio i.e. forced-beta of 9.

Q1 is either a standard high-voltage BJT device operating inreverse-beta mode, or an optimal transistor structure such as the B2device. SWA, SWB can be BJT, Mosfet or Jfet or other low-voltage typedevices.

The node VSS is a convenient internal circuit node relative to whichother voltages are to be measured and can be one terminal of anauto-generated power supply for internal control circuits, the otherbeing VDD.

We assume that the start-up process is complete, the D2-device isforward biased by the circuit current and Inductor current is flowing atall times in L1 i.e. continuous conduction mode (discontinuous mode isalso possible) and this current steered by switches SWA or SWB ontoeither the CE1 terminal (i.e. VSS) or the Base terminal of thetransistor. The inductor path is in the only DC conduction path throughthe ‘diode’ so it is generally apparent that its current must equal, onaverage, the load circuit current.

During phase A where SWA is on, the inductor current in L1 builds upwith a delta-I component at a rate set by V(C1) voltage (which is oforder 70 mV)/L1, and will exceed the load current by the end of Phase Atime. FIG. 38C has some illustrative waveforms.

In phase B, where SWA is switched off and SWB is switched on, theinductor current generally continues but voltage will instantaneouslyboost to a voltage set by the BASE of Q1—typically 0.75V (plus a littlefor the voltage drop of SWB). The inductor current droops during thistime at a rate set by (VBE−V(C1))/L1, dropping below the load currentvalue. Transistor Q1 has high, current-controlled, stored chargerepresented by Cbase and slow turnoff speed such that thesediscontinuous BASE current pulses are averaged into an effective basecurrent within the transistor. It can be seen that there is no netcharge given to Cbase in a cycle where 1 A*450 nS=9 A*50 nS.

The time-slicing of the inductor current between SWA and SWB paths intoCE1 and BASE terminals respectively gives a precise ‘Forced-Beta’ ratioto Q1 of t(PhaseA):t(PhaseB) This ratio is maintained over all loadcurrents without need for control system intervention and thereby keepsthe transistor optimally driven.

It was noted that the inductor sees a ripple current exceeding thendropping below the load-current value during the switching cycle. Sincethe load current value doesn't change over the short switching cycle, itfollows that the L1 ripple current must be balanced by an equal ripplecurrent to/from C1. C1 is therefore critical and is scaled up inproportion to expected inductor ripple current (which itself isinversely proportional to the chosen L1 inductor value), scaled up toreduce voltage ripple at the external terminals of the D2-device, andscaled down as the operating frequency increases.

Typical values for a 10 A ‘diode’ switching at 1 MHz are L1=10 nH, C1=47uF, Cbase=22 uF (using standard BJT). Cbase is generally not needed whenusing a B2 type device typically operating at 100 A/cm2 current densityand a forced-beta of 9. It is important to note that C1 only has to berated at a few volts and can be a very small ceramic type capacitor.Inductor L1 is of such small value that it can occur naturally from‘parasitic’ bond-wire inductance or PCB trace and need not useadditional magnetic materials.

Effective Vf of the ‘Diode’:

The object of this embodiment is to make a diode with a much lowerforward voltage than a standard PN junction diode.

Ignoring the small losses from parasitic resistances, the effectiveforward voltage of the ‘diode’ are made up of two components. 1) theaverage voltage over C1 plus 2) VCE(sat) of Q1.

The first component can be calculated from the voltage boost ratiobetween V(C1) and VBE of Q1. The voltage boost ratio is (t(PhaseA)/t(Phase B))+1, because of what is essentially a boost converteroperating with L1, SWA, SWB and the +1 resulting from the fact thatV(C1) is additive with the inductor boost voltage for driving VBE. SinceVBE is typically 0.75V, and if the t(Phase A):t(Phase B) ratio [i.e.forced-beta] is chosen to be 9:1 then C1 will settle to 75 mV when theconverter reaches equilibrium i.e. no net change of inductor currentover a complete cycle. Adding the typical 50 mV VCE(sat) of Q1 gives atotal ‘diode’ Vf drop of 0.125V in this example.

Vf can be reduced further by running with higher forced-beta settings(requires higher Beta of the transistor) and lower VCE(sat) which comesfrom using larger transistor operating at lower current density.Thinned-wafer B2 devices designed for 650V breakdown exhibit Betas ofbetween 30 and 10.

Reverse Blocking Characteristic:

In reverse-bias, and with C1 discharged, Q1 acts as a high voltage ‘off’NPN transistor with its Base biased to its Emitter by the leakageresistance of the active circuitry, or extra ‘pull-down’ resistanceadded if required. The reverse breakdown voltage is fully thatdetermined by the design of Q1.

Self-Oscillating Version:

FIG. 37 D,E,F describe a working system which uses self-sustainingoscillation provided by the addition of coupled-inductor L2 and twolow-voltage discrete BJT devices Q2, Q3 to implement SWA and SWB. An LCoscillator is formed from the coupling of L1 and L2 where a typical 9:1turns ratio can in principle produce a 0.75V Vbe to turn on Q2 base whenV(C1) is only 75 mV. This circuit has positive feedback and oscillatesnaturally between Phase A and Phase B operation as voltage C1 ripples upand down at a rate set by L1. L3 here is used to reduce the forwardvoltage of Q3 which is acting as SWB and could equally be replaced byother rectifier means such as Schottky diode, PNP with grounded baseeither of which method would no longer require L3. If VBE of Q2 issimilar to VBE of Q1 then the PWM duty cycle is similar to the L2:L1turns ratio.

Integrated SWA, SWB and Power transistor for self-oscillation version.

FIG. 37C is a cross section of a device which incorporates Q1 as avertical high voltage power device with Q2 and Q3 as low-voltagelateral/vertical devices giving a monolithic semiconductor solution.

NMOS Vs. BJT for SWA, SWB

Using BJTs for SWA and SWB limits the ultimate forward voltage of theD2-device to around 200 mV because BJTs do not quite saturate down to 0Vas do MOS devices. Using NMOS devices, perhaps in another die to Q1, andusing more accurate pulse timing than can be achieved withself-oscillation can bring the total loss to around 100 mV at low cost.

Driving SWA and SWB from a more sophisticated control circuit than asimple oscillator has advantages for the ‘diode’ application, butparticularly for Transistor/Thyristor type applications and especiallyfor smart-power devices.

C2-Device

FIG. 38A shows the concept of a device called C2 with the C referring toCMOS processing which would be typically used although NMOS processcould be used for lower cost.

Although this could be built from two separate die—a CMOS controller ICmounted on top of or alongside an I2 switch, FIG. 38B,D have thepreferred embodiment where the I2 devices if formed underneath true CMOSstructures on a monolithic wafer. This means that all types of circuitsand IP blocks can be used to make a self-powered, smart-power transistorwith fully built-in driver electronics e.g. ADC, DAC, Flash Memory, RAM,Microprocessor. An isolated control interface can be added using an offor on-die transformer making an easy to use smart transistor.

SWA and SWB are NMOS devices which on a typical 0.18 u CMOS process havea specific on resistance of 0.5 mOhm*mm2. To handle 10 A, only 1 mm2 ofsilicon area is needed to give a negligible 5 millivolts of DC voltagedrop in SWA.

A digital controller drives the gates of SWA and SWB running from a VDDsupply of typically 1.8 volts and switching at around 2 MHz. VDD isgenerated readily using Phase C period shown on the waveforms of FIG.38C as the short spike on the L1 voltage trace. A VDD voltage regulatorfunction can be completed by control loop using on-chip components toadjust the time period or occurrence rate of the Phase C period.

Rdamp is the combination of real and parasitic resistance designed todampen oscillations as the PWM forced-beta ratio is changed.

Further Refinements to the Control Techniques:

First order BJT base current control is automatic—the base current isalways a fixed fraction of collector current over all possible collectorcurrent range—set precisely by the PWM ratio giving the Forced-beta.

It is possible to add a second-order loop which increases the PWM ratioin proportion to the measured collector current. This can account forthe roll-off in Beta with increasing collector current of sometransistor types. For high-speed low-loss turn off it has been foundbest to reduce the force-beta to a value below the sustaining value andonly shutting of base current when the VCE1CE2 voltage is seen to rise,rather than simply cutting off base current immediately.

Refinements Made Easier with CMOS Integration:

With multiple stripe construction and very fine granularity oftransistor formation a C2-device can contain vertical I2 transistorformations of different characteristics on the same die. Dramaticallydifferent characteristics result from changing the backside emitter frombeing opaque to being transparent to holes using multiple maskedimplants at the rear of the device. FIG. 38B shows a slow device besidea fast device.

NMOS devices SWB can be doubled-up to give independent Base_slow andBase_fast destinations for the base current. This technique will combinethe best of high-Beta, low VCE(sat) characteristic of the slow devicewith the low Eoff losses of the fast device by arranging for a two-stageturn-off where first the slow transistor is turned off, handoff ispassed over by first turning on the fast transistor before this isturned off to exploit it very low Eoff characteristic.

In practice there would be larger segregation distances between the fastand slow devices—ideally the carrier diffusion length to reducecross-coupling. Also, the extremes of fully opaque and fully transparentmight not be used.

Start-Up Operation:

The self-oscillating ‘diode’ of FIG. 37D starts up within 10 uS of beingdriven in the forward direction once Q3 and Q1 PN junction voltages areexceeded (1.5V approx). Then C1 is quickly charged to around 1V andoscillation begins strongly. This initially higher forward voltagepersists for such a short time relative to the 1000× longer 50 Hzhalf-cycle time that it can be ignored for mains rectificationapplication

C2-Device Start-Up and Standby:

The C2-device must operate at up to 100 KHz but an initial delay of evena few seconds from system power-up is acceptable. Using either thetransistor's inherent leakage current or deliberate leakage path asformed by high-voltage JFET structure can charge capacitor C1 from thistiny current. With the extremely low standby power consumption inherentin CMOS, V(C1) can ramp to 2.5V which will give an initial VDD of 1.8V(one diode drop—see FIG. 38A).

This represents a standby condition where the energy stored on C1 isenough to begin which the device is ready to come into operational modeas soon as an input signal is received. Operational mode is as describedearlier with Phase A, B, C and self-powering technique. Dropping backinto standby mode means the device is ready to turn on again wheneverrequired without delay.

Advanced Bridge Rectifier with Value Added Features:

FIG. 39 shows ideas for combining active diodes and together to producefirst a basic low-loss bridge rectifier then an more advanced versionwith several added features

1) Overvoltage protection for downstream components. In the version ofFIG. 39B, an on-chip comparator on the CMOS die detects a mainsovervoltage and where the controlled transistor is a T2 device (asdescribed above), it can be tuned off, protecting any downstreamelectronics including electrolytic capacitors and power transistors fromovervoltage. These components can then be rated only for the typically450V excursions rather than rated at 650V device for infrequent mainsover voltages. Either cost savings, performance improvement and/orreliability increases can be expected from lower voltage, less stresseddevices.2) Soft start

Normally an additional device such as NTC thermistor is used in mainsrectification applications to reduce inrush currents into theelectrolytic capacitors at the expense of wasted heat in the NTCthermistor after the inrush event. The active bridge T2 type devices caninstead give a pulse-width modulated gradual charging of theelectrolytic during power-up before switching to the normallow-forward-voltage mode.

3) Auxiliary power supply output

A low voltage auxiliary power supply can be tapped from the internal VDDsupply and passed out of the bridge-rectifier module for use by externalcircuits such as Switch Mode Power Supply Controller or Power FactorController. These pins can be seen in FIG. 39D and FIG. 39E but are notbeing used in the example given.

Cost Estimates/Cost of Ownership

Standard Bridge rectifiers are very inexpensive devices, costing around$0.50 for a 10 A 600V plastic packaged bridge rectifier. This splits70%:30% between the cost of the diodes and the assembly/packagingmaterials.

However, the 15 watts minimum it will lose as heat costs incurs andextra $1 to deploy the necessary heatsinking means. Additionally, thecost of ownership is around $15 per year from electricity cost to fuelthe 15 W of heat it generates [based on mean world electricity price of$0.19 per KwHr]

D2-device bridge rectifier. 10 A bridge-rectifier made up according toFIG. 3E has the following costs.

-   -   CMOS device. Assume $800 per 8″ 24-mask wafer and kerf losses        @1.5 mm×1.5 mm=$0.057×4=$0.23    -   B2 device. Assume $200 per 8″ two-mask process wafer @3.3 mm×3.3        mm=$0.064×4=$0.255    -   Packaging +50% —package will have profiled shape to aid heat        dissipation.    -   Testing/Yield losses+20%

Total=˜$0.87

The price is competitive and is actually cheaper than the standardbridge rectifier if the price of its heatsink is included. Initial costcomparisons however are mute since using a D2-device bridges will saveits owner $12.50 per year per bridge at 10 A of rectified current.Payback time is in the order of 2 weeks. It also adds more than 1% tothe efficiency figure of a 120 Vac mains operated device which is animportant marketing metric for appliance manufacturers

It will be appreciated that the invention can be described in thefollowing clauses:

1. A power switching semiconductor device of bipolar construction and ofPNP or NPN structure where minority carriers are injected from a baseterminal which is normally kept inoperative and out-of-circuit by adepletion region induced by adjacent diffusion of opposite polarity andhigher doping or by a junction transistor formed where there isencroachment of the adjacent diffusion or a pre-deposited junction.

2. A power switching semiconductor device according to clause 1 wherehigh reverse bias voltages are supported by a lightly doped drift regionwhich comprises the Base of the transistor.

3. A power switching semiconductor device according to any precedingclause where conduction occurs in two quadrants supporting operation asan AC switch.

4. A power switching semiconductor device according to any precedingclause which has two base connections, one on each side of the wafer, tosupport efficient AC switching gain.

5. A power switching semiconductor device according to any precedingclause which has one base connection on one side of the wafer which isdriven by a single DC base supply.

6. A power switching semiconductor device according to any precedingclause with microprocessor controlled buck regulation of the basevoltage/current drive and analogue to digital feedback information intothe microprocessor's algorithms.

7. A power switching semiconductor device of PNP or NPN constructionwhere minority carriers are injected from a base terminal which isnormally kept inoperative and out-of-circuit by a depletion regioninduced by adjacent diffusion of opposite polarity and higher dopingwhere minority carriers are injected on the upper part of thesemiconductor using a dedicated emitter terminal and where twocollectors displaced laterally are formed on the lower side of thesemiconductor to form the switching terminals.

8. A power switching semiconductor device according to any precedingclause where the energy used to power the base of the device is derivedfrom the conduction voltage drop of the device.

9. A power switching semiconductor device according to any precedingclause where emitter/collector regions are etched then subsequentlydiffused with dopant.

10. A power switching semiconductor device according to any precedingclause where Quasi-PNP (for overall NPN device) or NPN (for overall PNPconstruction)/Quasi JFET electrode structure is present.

11. A power switching semiconductor device according to any precedingclause where conduction is synchronised to mains voltage cycle waveformfor zero-crossing switching.

12. A power switching semiconductor device according to any precedingclause used to synchronously rectify mains AC power to reduce powerlosses compared to the Vf of a standard semiconductor diode.

13. A power switching semiconductor device according to any precedingclause where Polysilicon trench fill is used to form emitter/collectorsof high doping and/or thin inter-facial oxide feature.

14. A power switching semiconductor device according to any precedingclause where collector/emitters suffer reduced minority carrierinjection through means of Silicon Germanium or other electric fieldgrading technique.

15. A power switching semiconductor device according to any precedingclause forming 3d or stacked devices to give higher power ability and/orhigher sensitivity and lower conduction losses.

16. A power switching semiconductor device according to any precedingclause where the stacked structure is used to increase the surface areaof the finished article to obviate the need or a dedicated heat-sink.

17. A power switching semiconductor device according to any precedingclause which incorporates a charge-control model of bipolar diffusioncurrent transport within its algorithms.

18. A power switching semiconductor device according to any precedingclause which is self-powered from the load using an auxiliary transistortap circuit switching on around the zero-crossing times.

19. A power switching semiconductor device according to any precedingclause where the finished article is calibrated and coefficients storedin non-volatile memory mounted with the transistor.

20. A power switching semiconductor device according to any precedingclause using a recessed BASE contact (auxiliary transistor emitter) sothat the a CE electrode is prominent allowing 3D device stacking withinterspaced conductor sheets for heat extraction and electricalconduction.

21. A power switching semiconductor device according to any precedingclause using a transformer coupling arrangement of LF power waveform andRF data waveform allowing for data networking and isolated power betweenan number of intelligent nodes.

22. A power switching semiconductor device according to any precedingclause where the required control system can be powered using theinbuilt boost converter acting on the energy available from the throughcurrent of the main semiconductor switch.

23. A power switching semiconductor module according to any precedingclause acting in the overall capacity as a two terminal fuse for acircuit to be protected.

24. A power switching semiconductor with a current-mode base controlwhere resistive DAC or DACs are used to control current of the overallor individual bases of the power semiconductor according to a controlprogram reactive to the measured operating conditions of the device.

25. An array of power switching semiconductors according to anyproceeding clause combined with a multichannel control circuit to effecta matrix converting ac to ac power converter.

26. A power switching semiconductor according to any preceding clausewhere a buffer layer is inserted of opposite doping under the CE2 orCollector terminal and has doping around the standard concentrationranges typical for punch-through or filed-stop control layers used inIGBTs or Diodes.

27. A power switch driver comprising a coupled dual PWM system where thelatter is a high-frequency buck-mode converter using inductor and theformer is a standard PWM channel and where the cross modulation of thetwo PWMs across a base drive inductance affords a continuous dynamiccurrent control of a power switch with a high speed on and off ability.

28. A coupled dual PWM driver where PWM2 frequency is such that the offtime is of similar order or not much longer than the minority carrierlifetime ensuring that conductivity of the switched transistor remainsapproximately constant in the PWM2 period.

29. Multiple, independently programmable, copies of the dual PWM drivereach driving a separate finger or power device die to provide theability to control current and timing independently on each region of adie or multiple die.

30. A coupled multi-channel driver where multiple phase-offset PWM2 typedrivers each driving an inductor with a common point on the baseterminal of a transistor and a single PWM1 channel to drive standarddevices such as silicon-carbide NPN transistor.

31. A coupled multi-channel driver according to above where multiphaseoffsets create a higher effective PWM2 frequency to match the reducedminority-carrier lifetime of high speed transistors such assilicon-carbide.

32. An intelligent power-transistor driver with multi-channel outputs asdescribed in previous clauses, which applies during operationpre-programmed co-efficient which had been determined aftermanufacturing to make a device with the overall uniform device voltage,current and temporal characteristics of a single device.

33. An intelligent power-transistor driver with multi-channel outputs asdescribed in previous clauses with an adaptive circuit to respond to anexternal PWM and by applying known delay coefficients and calibrationvalues known in advance to create the same PWM on/off ratios at theoutput transistor albeit with a slight overall delay.

34. An intelligent low leakage relay circuit of construction accordingto any preceding clause where a second switch shunts any leakage currentaround the load during switch off and optionally a third transistordisconnects to achieve Pico-ampere level leakage currents into the load.

35. An AC switchable power transistor and driver combination accordingto any preceding clause where a virtual diode action is formed by thedriver being commanded or detecting the need for reverse conduction andturning on the transistor accordingly.

36. A minority-carrier switching transistor whose base terminal andinternal structure current rating equal to that of the collector oremitter current rating so that it yields a reverse free-wheeling diodefunction of equal current rating to the forward rating and when the baseis actively clamped to ground using typically an NMOS FET from thedriver or when the base current is produced by the driver to effectreverse bias switch-on transistor action and to further reduce thevoltage of the C to E path below the normal Vdiode drop.

37. A High-side DC switch or multiple high side DC switches comprising atransistor according to any preceding clause and a single ormultichannel driver according to any preceding clause

Although the invention has been described in terms of preferredembodiments as set forth above, it should be understood that theseembodiments are illustrative only and that the claims are not limited tothose embodiments. Those skilled in the art will be able to makemodifications and alternatives in view of the disclosure which arecontemplated as falling within the scope of the appended claims. Eachfeature disclosed or illustrated in the present specification may beincorporated in the invention, whether alone or in any appropriatecombination with any other feature disclosed or illustrated herein.

1.-88. (canceled)
 89. A bi-directional bipolar junction transistor (BJT)structure, comprising: a base region of a first conductivity type,wherein said base region constitutes a drift region of said structure;first and second collector/emitter (CE) regions, each of a secondconductivity type adjacent opposite ends of said base region; whereinsaid base region is lightly doped relative to said collector/emitterregions; the structure further comprising: a base connection to saidbase region, wherein said base connection is within or adjacent to saidfirst collector/emitter region.
 90. A bi-directional BJT structure asclaimed in claim 89, wherein said first and second collector/emitterregions and said base region define a bi-directional BJT, and wherein aconnection to said base region of said bi-directional BJT via a secondtransistor having a first input/output (I/O) terminal connected to saidbase connection, and a control connection coupled to said first CEregion.
 91. A bi-directional BJT structure as claimed in claim 90,wherein said second transistor is a junction gate field-effecttransistor (JFET), wherein said base connection and base region aresource/drain connections of said JFET, wherein said control connectionis a gate terminal of said JFET, wherein said base connection isadjacent said first CE region, and wherein a channel region of said JFETis between said base connection and said base region.
 92. Abi-directional BJT as claimed in claim 90, wherein said secondtransistor is a BJT, wherein said first I/O terminal has said firstconductivity type, wherein said base connection is within said first CEregion, and wherein said control connection of said BJT is a base isformed by a portion of said first CE region.
 93. A bi-directional BJTstructure as claimed in claim 89, wherein said base region is wider in adirection between said ends of said base region than each of saidcollector/emitter regions, and wherein a current carrying capability ofa connection path between said base connection and said second CE regionis less than a current carrying capability of a connection path betweensaid first and second CE regions.
 94. A bi-directional BJT structure asclaimed in claim 89, wherein a forward conduction path from said baseregion to said second CE region is driven by a voltage on said baseregion relative to said second CE region, and wherein, when said forwardconduction path is present, a forward conduction path between said baseconnection and said base region includes a depleted portion of said baseregion.
 95. A bi-directional BJT structure as claimed in claim 89,wherein, when no voltage is applied to any terminals, the structure isin an off-state so as to form depletion regions between said first CEregion and base region and between said second CE region and baseregion.
 96. A bi-directional BJT structure as claimed in claim 89,wherein, when a positive voltage is applied to said second CE region andno voltage is applied to the first CE region and the base connection,the structure is in an off-state so as to form a depletion regionbetween said second CE region and base region.
 97. A bi-directional BJTstructure as claimed in claim 89, wherein, when a negative voltage isapplied to said second CE region and no voltage is applied to the firstCE region and the base connection, the structure is in an off-state soas to form a depletion region between said first CE region and baseregion.
 98. A bi-directional BJT structure as claimed in claim 89,wherein, when a first positive voltage is applied to said second CEregion, a second positive voltage being applied to the base connectionand no voltage is applied to the first CE region, the structure is in anon-state in which majority carriers from the first CE region flowthrough the base region towards the second CE region, and minoritycarriers from the base connection are injected into the base region, theminority carriers being recombined with the majority carriers in aregion adjacent the first CE region.
 99. A bi-directional BJT structureas claimed in claim 89, wherein, when a negative voltage is applied tosaid second CE region, a positive voltage being applied to the baseconnection and no voltage is applied to the first CE region, thestructure is in an on-state in which majority carriers from the secondCE region flow through the base region towards the first CE region, andminority carriers from the base connection are injected into the baseregion flowing towards the second CE region, the minority carriers beingrecombined with the majority carriers in a region adjacent the second CEregion.
 100. A bi-directional BJT structure as claimed in claim 89,wherein the first conductivity type comprises a p-type doping polarityand the second conductivity type comprises an n-type doping polarity;and/or wherein said structure is non-latching and switches off aconnection between said first and second CE regions on removal of avoltage from said base connection; and/or wherein said base connectionis recessed into a surface of said structure; and/or wherein said baseconnection is an ohmic connection comprising a region of said firstconductivity type, and wherein said base region is of ohmic type,wherein optionally said ohmic base region is configured to drivetransistor comprising the first CE region, base region and second CEregion into a saturation region during current conduction.
 101. Abi-directional BJT structure as claimed in claim 89, wherein said deviceis a vertical device.
 102. A bi-directional BJT structure as claimed inclaim 89, wherein said structure is a lateral structure.
 103. Abi-directional BJT structure as claimed in claim 89, wherein the secondCE region comprises two separation portions laterally disposed to oneanother and wherein each separate portion forms a switching terminal.104. A bipolar junction transistor (BJT) structure, comprising: a baseregion of a first conductivity type, wherein said base regionconstitutes a drift region of said structure; first and secondcollector/emitter (CE) regions, each of a second conductivity typeadjacent opposite ends of said base region; wherein said base region islightly doped relative to said collector/emitter regions; the structurefurther comprising: a base connection to said base region, wherein saidbase connection is within or adjacent to said first collector/emitterregion and a buried layer of the second conductivity type disposedbetween the second CE region and the base region.
 105. A BJT structureas claimed in claim 104, wherein the structure is configured to operatein a DC application.
 106. A bipolar junction transistor (BJT) structure,comprising: a base region of a first conductivity type, wherein saidbase region constitutes a drift region of said structure, the driftregion being a reverse voltage sustaining region; a collector region ofa second conductivity type; an emitter of a second conductivity type,the collector and emitter being adjacent opposite ends of said baseregion; wherein said base region is lightly doped relative to saidcollector and emitter regions; the structure further comprising: a baseconnection region of the first conductivity type formed adjacent to saidemitter region and a field stop layer of the first conductivity typeformed between the emitter region and the base region, the baseconnection being within the field stop layer.
 107. A BJT structure asclaimed in claim 106, wherein the doping concentration of the field stoplayer is less than that of the base connection; and/or wherein thethickness of the field stop layer is more than that of the baseconnection.
 108. A BJT structure as claimed in claim 106, beingconfigured such that a diode is formed between the collector and baseregion, wherein the diode is optionally configured to operate as areverse conducting diode when driven by a driver circuit.